The debate still goes on as to which are better, valves or transistors. We don’t intend to get involved in that argument here. But if you can’t make your mind up, you should try out this simple amplifier. This amplifier uses a valve as a pre-amplifier and a MOSFET in the output stage. The strong negative feedback makes the frequency response as flat as a pancake. In the prototype of the amplifier we’ve also tried a few alternative components.


For example, the BUZ11 can be replaced by an IRFZ34N and an ECC83 can be used instead of the ECC88. In that case the anode voltage should be reduced slightly to 155 V. The ECC83 (or its US equivalent the 12AX7) requires 2 x 6.3 V for the filament supply and there is no screen between the two triodes, normally connected to pin 9. This pin is now connected to the common of the two filaments.

The filaments are connected to ground via R5. If you’re keeping an eye on the quality, you should at least use MKT types for coupling capacitors C1, C4 and C7. Better still are MKP capacitors. For C8 you should have a look at Panasonic’s range of audio grade electrolytics. P1 is used to set the amount of negative feedback. The larger the negative feedback is, the flatter the frequency response will be, but the smaller the overall gain becomes.

Circuit diagram:
Simple Hybrid Audio Amplifier Circuit Diagram

With P2 you can set the quiescent current through T2. We have chosen a fairly high current of 1.3 A, making the output stage work in Class A mode. This does generate a relatively large amount of heat, so you should use a large heatsink for T2 with a thermal coefficient of 1 K/W or better. For L1 we connected two secondary windings in series from a 2x18V/225 VA toroidal transformer.

The resulting inductance of 150 mH was quite a bit more than the recommended 50 mH. However, with an output power of 1 W the amplifier had difficulty reproducing signals below 160 Hz. The distortion rose to as much as 9% for a signal of 20 Hz at 100 mW. To properly reproduce low-frequency signals the amplifier needs a much larger coil with an iron core and an air gap. This prevents the core from saturating when a large DC current flows through the coil.

Parts layout:

Such a core may be found in obsolete equipment, such as old video recorders. A suitable core consists of welded E and I sections. These transformers can be converted to the required inductor as follows: cut through the welding, remove the windings, add 250 to 300 windings of 0.8 mm enamelled copper wire, firmly fix the E and I sections back together with a piece of paper in between as isolation.

The concepts used in this circuit lend themselves very well to some experimentation. The number of supply voltages can be a bit of a problem to start with. For this reason we have designed a power supply especially for use with this amplifier (Quad power supply for hybrid amp). This can of course just as easily be used with other amplifiers. The supply uses a cascade stage to output an unstabilised voltage of 170 V for the SRPP (single rail push pull) stage (V1).

PCB layout:

During initial measurements we found that the ripple on this supply was responsible for a severe hum at the output of the amplifier. To get round this problem we designed a separate voltage regulator (High-voltage regulator with short circuit protection), which can cope with these high voltages. If you use a separate transformer for the filament supply you can try and see if the circuit works without R5. During the testing we used a DC voltage for the filament supply. Although you may not suspect it from the test measurements (see table), this amplifier doesn’t sound bad.

In fact, it is easily better than many consumer amplifiers. The output power is fairly limited, but is still enough to let your neighbours enjoy the music as well. It is possible to make the amplifier more powerful, in which case we recommend that you use more than one MOSFET in the output stage. The inductor also needs to be made beefier. Since this is a Class A amplifier, the supply needs to be able to output the required current, which becomes much greater at higher output powers. The efficiency of the amplifier is a bit over 30%.
Author: Frans Janssens - Copyright: Elektor Electronics

Simple Hybrid Audio Amplifier

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The debate still goes on as to which are better, valves or transistors. We don’t intend to get involved in that argument here. But if you can’t make your mind up, you should try out this simple amplifier. This amplifier uses a valve as a pre-amplifier and a MOSFET in the output stage. The strong negative feedback makes the frequency response as flat as a pancake. In the prototype of the amplifier we’ve also tried a few alternative components.


For example, the BUZ11 can be replaced by an IRFZ34N and an ECC83 can be used instead of the ECC88. In that case the anode voltage should be reduced slightly to 155 V. The ECC83 (or its US equivalent the 12AX7) requires 2 x 6.3 V for the filament supply and there is no screen between the two triodes, normally connected to pin 9. This pin is now connected to the common of the two filaments.

The filaments are connected to ground via R5. If you’re keeping an eye on the quality, you should at least use MKT types for coupling capacitors C1, C4 and C7. Better still are MKP capacitors. For C8 you should have a look at Panasonic’s range of audio grade electrolytics. P1 is used to set the amount of negative feedback. The larger the negative feedback is, the flatter the frequency response will be, but the smaller the overall gain becomes.

Circuit diagram:
Simple Hybrid Audio Amplifier Circuit Diagram

With P2 you can set the quiescent current through T2. We have chosen a fairly high current of 1.3 A, making the output stage work in Class A mode. This does generate a relatively large amount of heat, so you should use a large heatsink for T2 with a thermal coefficient of 1 K/W or better. For L1 we connected two secondary windings in series from a 2x18V/225 VA toroidal transformer.

The resulting inductance of 150 mH was quite a bit more than the recommended 50 mH. However, with an output power of 1 W the amplifier had difficulty reproducing signals below 160 Hz. The distortion rose to as much as 9% for a signal of 20 Hz at 100 mW. To properly reproduce low-frequency signals the amplifier needs a much larger coil with an iron core and an air gap. This prevents the core from saturating when a large DC current flows through the coil.

Parts layout:

Such a core may be found in obsolete equipment, such as old video recorders. A suitable core consists of welded E and I sections. These transformers can be converted to the required inductor as follows: cut through the welding, remove the windings, add 250 to 300 windings of 0.8 mm enamelled copper wire, firmly fix the E and I sections back together with a piece of paper in between as isolation.

The concepts used in this circuit lend themselves very well to some experimentation. The number of supply voltages can be a bit of a problem to start with. For this reason we have designed a power supply especially for use with this amplifier (Quad power supply for hybrid amp). This can of course just as easily be used with other amplifiers. The supply uses a cascade stage to output an unstabilised voltage of 170 V for the SRPP (single rail push pull) stage (V1).

PCB layout:

During initial measurements we found that the ripple on this supply was responsible for a severe hum at the output of the amplifier. To get round this problem we designed a separate voltage regulator (High-voltage regulator with short circuit protection), which can cope with these high voltages. If you use a separate transformer for the filament supply you can try and see if the circuit works without R5. During the testing we used a DC voltage for the filament supply. Although you may not suspect it from the test measurements (see table), this amplifier doesn’t sound bad.

In fact, it is easily better than many consumer amplifiers. The output power is fairly limited, but is still enough to let your neighbours enjoy the music as well. It is possible to make the amplifier more powerful, in which case we recommend that you use more than one MOSFET in the output stage. The inductor also needs to be made beefier. Since this is a Class A amplifier, the supply needs to be able to output the required current, which becomes much greater at higher output powers. The efficiency of the amplifier is a bit over 30%.
Author: Frans Janssens - Copyright: Elektor Electronics
Many modern motherboards are equipped with an infrared data interface compliant with the IrDA standard, but this interface not very often used. However, it is not difficult to build a data transmission module and connect it to the corresponding header. As can readily be seen from the schematic diagram, this doesn’t exactly involve a large array of ICs. This is because transceiver ICs are available for the IrDA standard, so only a few passive components have to be added to obtain an operational circuit. The author has successfully built this circuit many times using the TFDU5102 from Vishay Semiconductors (formerly Telefunken). If this IrDA transceiver is no longer available (it has been officially discontinued), the largely pin- and function-compatible TFDU6102 can be used without any problems.


IrDA Interface Circuit Diagram

This IC is faster and meets the latest IrDA specification. The TFDU6102 low-power receiver IC supports IrDA at data rates up to 4 Mbit/s (FIR), HP-SIR, Sharp ASK, and carrier-based remote control modes up to 2 MHz. The IC contains a photodiode, an infrared emitter and CMOS control logic. The IC also has internal protection against electromagnetic immissions and emissions, so no external screening is necessary. The IC works with a supply voltage of 2.7–5.5 V, so it is suitable for use in desktop PCs, notebooks, palmtops, and PDAs. It is also used in digital still and video cameras, printers, fax machines, copiers, projectors, and many other types of equipment.



The author has designed a printed circuit board for the IrDA module that is only 20 × 20 mm square. Of course, this means that all of the components are SMD types. The TFDU6102 in the ‘babyface’ package is available in upright and flat versions. Here the upright version (suffix ‘TR3’) is used. Thanks to its small size, the assembled circuit board can easily be placed behind a drive bay cover or the like. It is connected to the motherboard by a five-way flat cable. The pin assignments for header X1 must match the mating connector on the motherboard. After you have fitted the module, you may have to edit the BIOS settings to activate the UART for IrDA operation. These settings enable the (Windows) operating system to boot the new device and automatically install it. You may have to briefly insert the Windows CD to modify the settings. There is an abundance of free programs on the Internet that use the IrDA interface.

Resistors:
R1 = 7Ω5 (shape 1210)
R2 = 47 Ω (shape 1206)
R3 = 100 k (shape 1206)
Capacitors:
C1 = 100nF (shape 1206)
C2 = 4µF7 (shape 1210)
Semiconductors:
IC1 = TFDU6102TR3 (Vishay) (Farnell)
Miscellaneous:
X1 = 5-way SIL pinheader
Author: A. Bitzer
Copyright: Elektor Electronics

IrDA Interface

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Many modern motherboards are equipped with an infrared data interface compliant with the IrDA standard, but this interface not very often used. However, it is not difficult to build a data transmission module and connect it to the corresponding header. As can readily be seen from the schematic diagram, this doesn’t exactly involve a large array of ICs. This is because transceiver ICs are available for the IrDA standard, so only a few passive components have to be added to obtain an operational circuit. The author has successfully built this circuit many times using the TFDU5102 from Vishay Semiconductors (formerly Telefunken). If this IrDA transceiver is no longer available (it has been officially discontinued), the largely pin- and function-compatible TFDU6102 can be used without any problems.


IrDA Interface Circuit Diagram

This IC is faster and meets the latest IrDA specification. The TFDU6102 low-power receiver IC supports IrDA at data rates up to 4 Mbit/s (FIR), HP-SIR, Sharp ASK, and carrier-based remote control modes up to 2 MHz. The IC contains a photodiode, an infrared emitter and CMOS control logic. The IC also has internal protection against electromagnetic immissions and emissions, so no external screening is necessary. The IC works with a supply voltage of 2.7–5.5 V, so it is suitable for use in desktop PCs, notebooks, palmtops, and PDAs. It is also used in digital still and video cameras, printers, fax machines, copiers, projectors, and many other types of equipment.



The author has designed a printed circuit board for the IrDA module that is only 20 × 20 mm square. Of course, this means that all of the components are SMD types. The TFDU6102 in the ‘babyface’ package is available in upright and flat versions. Here the upright version (suffix ‘TR3’) is used. Thanks to its small size, the assembled circuit board can easily be placed behind a drive bay cover or the like. It is connected to the motherboard by a five-way flat cable. The pin assignments for header X1 must match the mating connector on the motherboard. After you have fitted the module, you may have to edit the BIOS settings to activate the UART for IrDA operation. These settings enable the (Windows) operating system to boot the new device and automatically install it. You may have to briefly insert the Windows CD to modify the settings. There is an abundance of free programs on the Internet that use the IrDA interface.

Resistors:
R1 = 7Ω5 (shape 1210)
R2 = 47 Ω (shape 1206)
R3 = 100 k (shape 1206)
Capacitors:
C1 = 100nF (shape 1206)
C2 = 4µF7 (shape 1210)
Semiconductors:
IC1 = TFDU6102TR3 (Vishay) (Farnell)
Miscellaneous:
X1 = 5-way SIL pinheader
Author: A. Bitzer
Copyright: Elektor Electronics
The National Semiconductor LMV225 is a linear RF power meter IC in an SMD package. It can be used over the frequency range of 450 MHz to 2000 MHz and requires only four external components. The input coupling capacitor isolates the DC voltage of the IC from the input signal. The 10-k? resistor enables or disables the IC according to the DC voltage present at the input pin. If it is higher than 1.8 V, the detector is enabled and draws a current of around 5–8 mA. If the voltage on pin A1 is less than 0.8 V, the IC enters the shutdown mode and draws a current of only a few microampères. The LMV225 can be switched between the active and shutdown states using a logic-level signal if the signal is connected to the signal via the 10-kR resistor.


Circuit diagram:

Linear RF Power Meter Circuit Diagram

The supply voltage, which can lie between +2.7 V und +5.5 V, is filtered by a 100nF capacitor that diverts residual RF signals to ground. Finally, there is an output capacitor that forms a low-pass filter in combination with the internal circuitry of the LMV225. If this capacitor has a value of 1 nF, the corner frequency of this low-pass filter is approximately 8 kHz. The corner frequency can be calculated using the formula fc = 1 ÷ (2 p COUT Ro) where Ro is the internal output impedance (19.8 k?). The output low-pass filter determines which AM modulation components are passed by the detector.


The output, which has a relatively high impedance, provides an output voltage that is proportional to the signal power, with a slope of 40 mV/dB. The output is 2.0 V at 9 dBm and 0.4 V at –40 dBm. A level of 0 dBm corresponds to a power of 1 mW in 50 R. For a sinusoidal wave-form, this is equivalent to an effective voltage of 224 mV. For modulated signals, the relationship between power and voltage is generally different. The table shows several examples of power levels and voltages for sinusoidal signals. The input impedance of the LMV225 detector is around 50 R to provide a good match to the characteristic impedance commonly used in RF circuits.

The data sheet for the LMV225 shows how the 40-dB measurement range can be shifted to a higher power level using a series input resistor. The LMV225 was originally designed for use in mobile telephones, so it comes in a tiny SMD package with dimensions of only around 1 × 1 mm with four solder bumps (similar to a ball-grid array package). The connections are labelled A1, A2, B1 and B1, like the elements of a matrix. The corner next to A1 is bevelled.
Author: Gregor Kleine
Copyright: Elektor Electronics

Linear RF Power Meter

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The National Semiconductor LMV225 is a linear RF power meter IC in an SMD package. It can be used over the frequency range of 450 MHz to 2000 MHz and requires only four external components. The input coupling capacitor isolates the DC voltage of the IC from the input signal. The 10-k? resistor enables or disables the IC according to the DC voltage present at the input pin. If it is higher than 1.8 V, the detector is enabled and draws a current of around 5–8 mA. If the voltage on pin A1 is less than 0.8 V, the IC enters the shutdown mode and draws a current of only a few microampères. The LMV225 can be switched between the active and shutdown states using a logic-level signal if the signal is connected to the signal via the 10-kR resistor.


Circuit diagram:

Linear RF Power Meter Circuit Diagram

The supply voltage, which can lie between +2.7 V und +5.5 V, is filtered by a 100nF capacitor that diverts residual RF signals to ground. Finally, there is an output capacitor that forms a low-pass filter in combination with the internal circuitry of the LMV225. If this capacitor has a value of 1 nF, the corner frequency of this low-pass filter is approximately 8 kHz. The corner frequency can be calculated using the formula fc = 1 ÷ (2 p COUT Ro) where Ro is the internal output impedance (19.8 k?). The output low-pass filter determines which AM modulation components are passed by the detector.


The output, which has a relatively high impedance, provides an output voltage that is proportional to the signal power, with a slope of 40 mV/dB. The output is 2.0 V at 9 dBm and 0.4 V at –40 dBm. A level of 0 dBm corresponds to a power of 1 mW in 50 R. For a sinusoidal wave-form, this is equivalent to an effective voltage of 224 mV. For modulated signals, the relationship between power and voltage is generally different. The table shows several examples of power levels and voltages for sinusoidal signals. The input impedance of the LMV225 detector is around 50 R to provide a good match to the characteristic impedance commonly used in RF circuits.

The data sheet for the LMV225 shows how the 40-dB measurement range can be shifted to a higher power level using a series input resistor. The LMV225 was originally designed for use in mobile telephones, so it comes in a tiny SMD package with dimensions of only around 1 × 1 mm with four solder bumps (similar to a ball-grid array package). The connections are labelled A1, A2, B1 and B1, like the elements of a matrix. The corner next to A1 is bevelled.
Author: Gregor Kleine
Copyright: Elektor Electronics
As electrolytic capacitors age, their internal resistance, also known as "equivalent series resistance" (ESR), gradually increases. This can eventually lead to equipment failure. Using this design, you can measure the ESR of suspect capacitors as well as other small resistances. Basically, the circuit generates a low-voltage 100kHz test signal, which is applied to the capacitor via a pair of probes. An op amp then amplifies the voltage dropped across the capacitor’s series resistance and this can be displayed on a standard multimeter. In more detail, inverter IC1d is configured as a 200kHz oscillator.

Its output drives a 4027 J-K flipflop, which divides the oscillator signal in half to ensure an equal mark/space ratio. Two elements of a 4066 quad bilateral switch (IC3c & IC3d) are alternately switched on by the complementary outputs of the J-K flipflop. One switch input (pin 11) is connected to +5V, whereas the other (pin 8) is connected to -5V. The outputs (pins 9 & 10) of these two switches are connected together, with the result being a ±5V 100kHz square wave. Series resistance is included to current-limit the signal before it is applied to the capacitor under test via a pair of test probes. Diodes D1 and D2 limit the signal swing and protect the 4066 outputs in case the capacitor is charged.

Circuit diagram:

ESR & Low Resistance Test Meter Circuit Diagram
A second pair of leads sense the signal developed across the probe tips. Once again, the signal is limited by diodes (D3 & D4) before begin applied to the remaining two inputs of the 4066 switch (pins 2 & 3 of IC3a & IC3b). These switches direct alternate half cycles to two 1μF capacitors, removing most of the AC component of the signal and providing a simple "sample and hold" mechanism. The 1μF capacitors charge to a DC level that is proportional to the test capacitor’s ESR. This is differentially amplified by op amp IC4 so that it can be displayed on a digital multimeter – 10Ω will be represented by 100mV, 1Ω by 10mV, etc. To calibrate the circuit, first adjust VR1 to obtain 100kHz at TP3.

Next, momentarily short the test probes together and adjust VR4 for 0mV at pin 6 of IC4. That done, set your meter to read milliamps and connect it between TP4 and the negative (-) DMM output. Apply -5V to TP2 and note the current flow, which should be around 2.1mA. Transfer the -5V from TP2 to TP1 and adjust VR2 until the same current (ignore sign) is obtained. Remove the -5V from TP1. Again, set to your meter to read volts and connect it to the DMM outputs. Apply the probes to a 10W resistor and adjust VR3 for a reading of 100mV. Finally, ensure that all capacitors to be tested are always fully discharged before connecting the probes.
Author: Len Cox - Copyright: Silicon Chip Electronics

ESR & Low Resistance Test Meter

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As electrolytic capacitors age, their internal resistance, also known as "equivalent series resistance" (ESR), gradually increases. This can eventually lead to equipment failure. Using this design, you can measure the ESR of suspect capacitors as well as other small resistances. Basically, the circuit generates a low-voltage 100kHz test signal, which is applied to the capacitor via a pair of probes. An op amp then amplifies the voltage dropped across the capacitor’s series resistance and this can be displayed on a standard multimeter. In more detail, inverter IC1d is configured as a 200kHz oscillator.

Its output drives a 4027 J-K flipflop, which divides the oscillator signal in half to ensure an equal mark/space ratio. Two elements of a 4066 quad bilateral switch (IC3c & IC3d) are alternately switched on by the complementary outputs of the J-K flipflop. One switch input (pin 11) is connected to +5V, whereas the other (pin 8) is connected to -5V. The outputs (pins 9 & 10) of these two switches are connected together, with the result being a ±5V 100kHz square wave. Series resistance is included to current-limit the signal before it is applied to the capacitor under test via a pair of test probes. Diodes D1 and D2 limit the signal swing and protect the 4066 outputs in case the capacitor is charged.

Circuit diagram:

ESR & Low Resistance Test Meter Circuit Diagram
A second pair of leads sense the signal developed across the probe tips. Once again, the signal is limited by diodes (D3 & D4) before begin applied to the remaining two inputs of the 4066 switch (pins 2 & 3 of IC3a & IC3b). These switches direct alternate half cycles to two 1μF capacitors, removing most of the AC component of the signal and providing a simple "sample and hold" mechanism. The 1μF capacitors charge to a DC level that is proportional to the test capacitor’s ESR. This is differentially amplified by op amp IC4 so that it can be displayed on a digital multimeter – 10Ω will be represented by 100mV, 1Ω by 10mV, etc. To calibrate the circuit, first adjust VR1 to obtain 100kHz at TP3.

Next, momentarily short the test probes together and adjust VR4 for 0mV at pin 6 of IC4. That done, set your meter to read milliamps and connect it between TP4 and the negative (-) DMM output. Apply -5V to TP2 and note the current flow, which should be around 2.1mA. Transfer the -5V from TP2 to TP1 and adjust VR2 until the same current (ignore sign) is obtained. Remove the -5V from TP1. Again, set to your meter to read volts and connect it to the DMM outputs. Apply the probes to a 10W resistor and adjust VR3 for a reading of 100mV. Finally, ensure that all capacitors to be tested are always fully discharged before connecting the probes.
Author: Len Cox - Copyright: Silicon Chip Electronics
In these times with viruses and other threats from the Internet it would be nice to have reassurance that the PC cannot be infected. That is why this circuit was designed. It makes it possible to install multiple hard disks inside the case of a PC, which are separated in such a way that viruses cannot move from one disk to another. In this case there are three drives installed, one for use of the Internet via ADSL, one for working with email and one for other applications.

If data from the Internet never arrives on the third disk, it is effectively protected against viruses. The solution outlined here has been in satisfactory use for a couple of years. There is an additional benefit: if there are ever any problems with the operation of the computer, then it is very easy to change to another hard disk to check if the problem manifests itself there as well. In this case, fault finding can be made much easier. The circuit operates by only switching over the power supply voltages (5 V and 12 V) of the hard disks. The hard disk is out of service without a power supply. This works without a problem with S-ATA disks.

Circuit diagram:
Hard Disk Switch Circuit Diagram

With IDE disks this only works with modern drives. There may only be a combination of hard disks on the relevant port and no CD-ROM, DVD-drive, CD-burner or something similar. The selection of the desired hard disk is done with a rotary switch. This has to be set to the correct position before the computer is switched on. When the power supply is turned on, one of three relays is driven via diode D1, D2 or D3. The relays are provided with a hold circuit via a second diode (D4, D5 and D6). In this way the selected relay remains energised as long as the power supply voltage is present.

After switching on, electrolytic capacitor C1 is charged via R1, so that the common contact of the rotary switch is quickly at 0 V. This prevents an accidental change of hard disk while the computer is in operation. The ADSL modem is powered from the PC. This power supply voltage is only present if hard disk number 2 is selected. This prevents the use of the Internet if one of the other disks is selected.
Author: Uwe Kardel - Copyright: Elektor Electronics Magazine

Hard Disk Switch

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In these times with viruses and other threats from the Internet it would be nice to have reassurance that the PC cannot be infected. That is why this circuit was designed. It makes it possible to install multiple hard disks inside the case of a PC, which are separated in such a way that viruses cannot move from one disk to another. In this case there are three drives installed, one for use of the Internet via ADSL, one for working with email and one for other applications.

If data from the Internet never arrives on the third disk, it is effectively protected against viruses. The solution outlined here has been in satisfactory use for a couple of years. There is an additional benefit: if there are ever any problems with the operation of the computer, then it is very easy to change to another hard disk to check if the problem manifests itself there as well. In this case, fault finding can be made much easier. The circuit operates by only switching over the power supply voltages (5 V and 12 V) of the hard disks. The hard disk is out of service without a power supply. This works without a problem with S-ATA disks.

Circuit diagram:
Hard Disk Switch Circuit Diagram

With IDE disks this only works with modern drives. There may only be a combination of hard disks on the relevant port and no CD-ROM, DVD-drive, CD-burner or something similar. The selection of the desired hard disk is done with a rotary switch. This has to be set to the correct position before the computer is switched on. When the power supply is turned on, one of three relays is driven via diode D1, D2 or D3. The relays are provided with a hold circuit via a second diode (D4, D5 and D6). In this way the selected relay remains energised as long as the power supply voltage is present.

After switching on, electrolytic capacitor C1 is charged via R1, so that the common contact of the rotary switch is quickly at 0 V. This prevents an accidental change of hard disk while the computer is in operation. The ADSL modem is powered from the PC. This power supply voltage is only present if hard disk number 2 is selected. This prevents the use of the Internet if one of the other disks is selected.
Author: Uwe Kardel - Copyright: Elektor Electronics Magazine
How often does it happen that you close down Windows and then forget to turn off the computer? This circuit does that automatically. After Windows is shut down there is a ‘click’ a second later and the PC is disconnected from the mains. Surprisingly enough, this switch fits in some older computer cases. If the circuit doesn’t fit then it will have to be housed in a separate enclosure. That is why a supply voltage of 5 V was selected. This voltage can be obtained from a USB port when the circuit has to be on the outside of the PC case.

It is best to solder the mains wires straight onto the switch and to insulate them with heat shrink sleeving. C8 is charged via D1. This is how the power supply voltage for IC1 is obtained. A square wave oscillator is built around IC1a, R1 and C9, which drives inverters IC1c to f. The frequency is about 50 kHz. The four inverters in parallel power the voltage multiplier, which has a multiplication of 3, and is built from C1 to C3 and D2 to D5. This is used to charge C5 to C7 to a voltage of about 9 V.

The generated voltage is clearly lower than the theoretical 3x4.8=14.4 V, because some voltage is lost across the PN-junctions of the diodes. C5 to C7 form the buffer that powers the coil of the switch when switching off. The capacitors charge up in about two seconds after switching on. The circuit is now ready for use. When Windows is closed down, the 5-V power supply voltage disappears. C4 is discharged via R2 and this results in a ‘0’ at the input of inverter IC1b. The output then becomes a ‘1’, which causes T1 to turn on.

Circuit diagram:
Computer Off Switch Circuit Diagram

A voltage is now applied to the coil in the mains switch and the power supply of the PC is turned off. T1 is a type BSS295 because the resistance of the coil is only 24R. When the PC is switched on, the circuit draws a peak current of about 200 mA, after which the current consumption drops to about 300 µA. The current when switching on could be higher because this is strongly dependent on the characteristics of the 5-V power supply and the supply rails in the PC. There isn’t much to say about the construction of the circuit itself.

The only things to take care with are the mains wires to the switch. The mains voltage may not appear at the connections to the coil. That is why there has to be a distance of at least 6 mm between the conductors that are connected to the mains and the conductors that are connected to the low-voltage part of the circuit.
Author: Uwe Kardel - Copyright: Elektor Electronics Magazine

Computer Off Switch

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How often does it happen that you close down Windows and then forget to turn off the computer? This circuit does that automatically. After Windows is shut down there is a ‘click’ a second later and the PC is disconnected from the mains. Surprisingly enough, this switch fits in some older computer cases. If the circuit doesn’t fit then it will have to be housed in a separate enclosure. That is why a supply voltage of 5 V was selected. This voltage can be obtained from a USB port when the circuit has to be on the outside of the PC case.

It is best to solder the mains wires straight onto the switch and to insulate them with heat shrink sleeving. C8 is charged via D1. This is how the power supply voltage for IC1 is obtained. A square wave oscillator is built around IC1a, R1 and C9, which drives inverters IC1c to f. The frequency is about 50 kHz. The four inverters in parallel power the voltage multiplier, which has a multiplication of 3, and is built from C1 to C3 and D2 to D5. This is used to charge C5 to C7 to a voltage of about 9 V.

The generated voltage is clearly lower than the theoretical 3x4.8=14.4 V, because some voltage is lost across the PN-junctions of the diodes. C5 to C7 form the buffer that powers the coil of the switch when switching off. The capacitors charge up in about two seconds after switching on. The circuit is now ready for use. When Windows is closed down, the 5-V power supply voltage disappears. C4 is discharged via R2 and this results in a ‘0’ at the input of inverter IC1b. The output then becomes a ‘1’, which causes T1 to turn on.

Circuit diagram:
Computer Off Switch Circuit Diagram

A voltage is now applied to the coil in the mains switch and the power supply of the PC is turned off. T1 is a type BSS295 because the resistance of the coil is only 24R. When the PC is switched on, the circuit draws a peak current of about 200 mA, after which the current consumption drops to about 300 µA. The current when switching on could be higher because this is strongly dependent on the characteristics of the 5-V power supply and the supply rails in the PC. There isn’t much to say about the construction of the circuit itself.

The only things to take care with are the mains wires to the switch. The mains voltage may not appear at the connections to the coil. That is why there has to be a distance of at least 6 mm between the conductors that are connected to the mains and the conductors that are connected to the low-voltage part of the circuit.
Author: Uwe Kardel - Copyright: Elektor Electronics Magazine

Charge your iPod without connecting it to a computer!

Using the USB port on your computer to charge your player’s batteries is not always practical. What if you do not have a computer available at the time or if you do not want to power up a computer just for charging? Or what if you are traveling? Chargers for Mobile Phones iPods and MP3 players are available but they are expensive and you need separate models for charging at home and in the car.

This charger can be used virtually anywhere. While we call the unit a charger, it really is nothing more than a 5V supply that has a USB outlet. The actual charging circuit is incorporated within the iPOD or MP3 player itself, which only requires a 5V supply. As well as charging, this supply can run USB-powered accessories such as reading lights, fans and chargers, particularly for mobile phones.

The supply is housed in a small plastic case with a DC input socket at one end and a USB type "A" outlet at the other end, for connecting to Mobile Phone, an iPod or MP3 player when charging. A LED shows when power is available at the USB socket. Maximum current output is 660mA, more than adequate to run any USB-powered accessory.

Pictures, PCB and Circuit Diagram:

Front View Of Mobile Phone and iPod Battery Charger Circuit


Bottom View Of Mobile Phone and iPod Battery Charger Circuit




PCB Layout Of Mobile Phone and iPod Battery Charger Circuit


Mobile Phone and iPod Battery Charger Circuit Diagram

Parts:
P1 = 1K
R1 = 1R-0.5W
R2 = 1R-0.5W
R3 = 1R-0.5W
R4 = 1K
R5 = 560R
R6 = 10R-0.5W
R7 = 470R
C1 = 470uF-25V
C2 = 100nF-63V
C3 = 470pF
C4 = 100uF-25V
D1 = 1N5404
D2 = 1N4001
D3 = 1N5819
D4 = 5.1V-1W Zener Diode
D5 = 5mm. Red LED
L1 = 220uH
S1 = USB 'A' Type Socket
SW1 = On/Off Switch
IC1 = MC34063A

Specifications:
Output voltage ----------------------5V
Output current ---------------------660mA maximum for 5V out
Input voltage range ------------------9.5V to 15V DC
Input current requirement ----------500mA for 9V in, 350mA for >12V input
Input current with output shorted--- 120mA at 9V in, 80mA at 15V in
Output ripple ------------------------14mV (from no load to 660mA)
Load regulation ----------------------25mV (from no load to 660mA)
Line regulation ----------------------20mV change at full load from 9 to 18V input
No load input current ----------------20mA

(The specification for the computer USB 2.0 port requires the USB port to deliver up to 500mA at an output voltage between 5.25V and 4.375V).

The circuit is based around an MC34063 switch mode regulator. This has high efficiency so that there is very little heat produced inside the box, even when delivering its maximum output current. The circuit is more complicated than if we used a 7805 3-terminal regulator but since the input voltage could be 15V DC or more, the voltage dissipation in such a regulator could be 5W or more at 500mA. and 5W is far too much for a 7805, even with quite a large heatsink. Credit for this circuit goes to SiliconChip, A wonderful electronics magazine.

Mobile Phone and iPod Battery Charger Circuit

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Charge your iPod without connecting it to a computer!

Using the USB port on your computer to charge your player’s batteries is not always practical. What if you do not have a computer available at the time or if you do not want to power up a computer just for charging? Or what if you are traveling? Chargers for Mobile Phones iPods and MP3 players are available but they are expensive and you need separate models for charging at home and in the car.

This charger can be used virtually anywhere. While we call the unit a charger, it really is nothing more than a 5V supply that has a USB outlet. The actual charging circuit is incorporated within the iPOD or MP3 player itself, which only requires a 5V supply. As well as charging, this supply can run USB-powered accessories such as reading lights, fans and chargers, particularly for mobile phones.

The supply is housed in a small plastic case with a DC input socket at one end and a USB type "A" outlet at the other end, for connecting to Mobile Phone, an iPod or MP3 player when charging. A LED shows when power is available at the USB socket. Maximum current output is 660mA, more than adequate to run any USB-powered accessory.

Pictures, PCB and Circuit Diagram:

Front View Of Mobile Phone and iPod Battery Charger Circuit


Bottom View Of Mobile Phone and iPod Battery Charger Circuit




PCB Layout Of Mobile Phone and iPod Battery Charger Circuit


Mobile Phone and iPod Battery Charger Circuit Diagram

Parts:
P1 = 1K
R1 = 1R-0.5W
R2 = 1R-0.5W
R3 = 1R-0.5W
R4 = 1K
R5 = 560R
R6 = 10R-0.5W
R7 = 470R
C1 = 470uF-25V
C2 = 100nF-63V
C3 = 470pF
C4 = 100uF-25V
D1 = 1N5404
D2 = 1N4001
D3 = 1N5819
D4 = 5.1V-1W Zener Diode
D5 = 5mm. Red LED
L1 = 220uH
S1 = USB 'A' Type Socket
SW1 = On/Off Switch
IC1 = MC34063A

Specifications:
Output voltage ----------------------5V
Output current ---------------------660mA maximum for 5V out
Input voltage range ------------------9.5V to 15V DC
Input current requirement ----------500mA for 9V in, 350mA for >12V input
Input current with output shorted--- 120mA at 9V in, 80mA at 15V in
Output ripple ------------------------14mV (from no load to 660mA)
Load regulation ----------------------25mV (from no load to 660mA)
Line regulation ----------------------20mV change at full load from 9 to 18V input
No load input current ----------------20mA

(The specification for the computer USB 2.0 port requires the USB port to deliver up to 500mA at an output voltage between 5.25V and 4.375V).

The circuit is based around an MC34063 switch mode regulator. This has high efficiency so that there is very little heat produced inside the box, even when delivering its maximum output current. The circuit is more complicated than if we used a 7805 3-terminal regulator but since the input voltage could be 15V DC or more, the voltage dissipation in such a regulator could be 5W or more at 500mA. and 5W is far too much for a 7805, even with quite a large heatsink. Credit for this circuit goes to SiliconChip, A wonderful electronics magazine.
Unplugging or re-connecting equipment to the serial COM or PS2 connector always gives problems if the PC is running. Even if you only need to swap a mouse or changeover from a graphics keyboard to a standard keyboard. The chances are that the connected equipment will not communicate with the PC, it will always be necessary to re-boot. If you are really unlucky you may have damaged the PC or the peripheral device. In order to switch equipment successfully it is necessary to follow a sequence. The clock and data lines need to be disconnected from the device before the power line is removed. And likewise the power line must be connected first to the new device before the clock and data lines are re-connected.

This sequence is also used by the USB connector but achieved rather more simply by using different length pins in the connector. The circuit shown here in Figure 1 performs the switching sequence electronically. The clock and data lines from the PC are connected via the N.C. contacts of relay RE2 through the bistable relay RE1 to connector K3. Pressing push-button S1 will activate relay RE2 thereby disconnecting the data and clock lines also while S1 is held down the semiconductor switch IC1B will be opened, allowing the voltage on C4 to charge up through R4. After approximately 0.2 s the voltage level on C4 will be high enough to switch on IC1A, this in turn will switch on T1 energizing one of the coils of the bistable relay RE1 and routing the clock, data and power to connector K2.


When S1 is released relay RE2 will switch the data and clock lines through to the PC via connector K1. It should be noted that the push-button must be pressed for about 0.5s otherwise the circuit will not operate correctly. Switching back over to connector K3 is achieved similarly by pressing S2. The current required to switch the relays is relatively large for the serial interface to cope with so the energy necessary is stored in two relatively large capacitors (C2 and C3) and these are charged through resistors R1 and R6 respectively. The disadvantage is that the circuit needs approximately 0.5 minute between switch-overs to ensure these capacitors have sufficient charge.

The current consumption of the entire circuit however is reduced to just a few milliamps. The PCB is designed to accept PS2 style connectors but if you are using an older PC that needs 9 pin sub D connectors then these will need to be connected to the PCB via flying leads. In this case the mouse driver software configures pin 9 as the clock, pin 1 as the data, pin 8 (CTS) as the voltage supply pin and pin 5 as earth.

Resistors:
R1 = 2kΩ2
R2 = 47kΩ
R3 = 10kΩ
R4 = 4kΩ7
R5 = 1kΩ
R6 = 1kΩ2
Capacitors:
C1 = 10µF 10V radial
C2 = 1000µF 10V radial
C3 = 2200µF 10V radial
C4 = 2µF2 10V radial
Semiconductors:
D1-D5 = 1N4148
T1 = BC547
IC1 = 4066 or 74HCT4066
Miscellaneous:
RE1 = bistable relay 4 c/o contacts
RE2 = monostable relay 2 c/o contacts
K1,K2,K3 = 6-way Mini-DIN socket (pins at 240°, PCB mount
S1,S2 = push-button (ITTD6-R)
Source: extremecircuits.net

Keyboard/Mouse Switch Unit

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Unplugging or re-connecting equipment to the serial COM or PS2 connector always gives problems if the PC is running. Even if you only need to swap a mouse or changeover from a graphics keyboard to a standard keyboard. The chances are that the connected equipment will not communicate with the PC, it will always be necessary to re-boot. If you are really unlucky you may have damaged the PC or the peripheral device. In order to switch equipment successfully it is necessary to follow a sequence. The clock and data lines need to be disconnected from the device before the power line is removed. And likewise the power line must be connected first to the new device before the clock and data lines are re-connected.

This sequence is also used by the USB connector but achieved rather more simply by using different length pins in the connector. The circuit shown here in Figure 1 performs the switching sequence electronically. The clock and data lines from the PC are connected via the N.C. contacts of relay RE2 through the bistable relay RE1 to connector K3. Pressing push-button S1 will activate relay RE2 thereby disconnecting the data and clock lines also while S1 is held down the semiconductor switch IC1B will be opened, allowing the voltage on C4 to charge up through R4. After approximately 0.2 s the voltage level on C4 will be high enough to switch on IC1A, this in turn will switch on T1 energizing one of the coils of the bistable relay RE1 and routing the clock, data and power to connector K2.


When S1 is released relay RE2 will switch the data and clock lines through to the PC via connector K1. It should be noted that the push-button must be pressed for about 0.5s otherwise the circuit will not operate correctly. Switching back over to connector K3 is achieved similarly by pressing S2. The current required to switch the relays is relatively large for the serial interface to cope with so the energy necessary is stored in two relatively large capacitors (C2 and C3) and these are charged through resistors R1 and R6 respectively. The disadvantage is that the circuit needs approximately 0.5 minute between switch-overs to ensure these capacitors have sufficient charge.

The current consumption of the entire circuit however is reduced to just a few milliamps. The PCB is designed to accept PS2 style connectors but if you are using an older PC that needs 9 pin sub D connectors then these will need to be connected to the PCB via flying leads. In this case the mouse driver software configures pin 9 as the clock, pin 1 as the data, pin 8 (CTS) as the voltage supply pin and pin 5 as earth.

Resistors:
R1 = 2kΩ2
R2 = 47kΩ
R3 = 10kΩ
R4 = 4kΩ7
R5 = 1kΩ
R6 = 1kΩ2
Capacitors:
C1 = 10µF 10V radial
C2 = 1000µF 10V radial
C3 = 2200µF 10V radial
C4 = 2µF2 10V radial
Semiconductors:
D1-D5 = 1N4148
T1 = BC547
IC1 = 4066 or 74HCT4066
Miscellaneous:
RE1 = bistable relay 4 c/o contacts
RE2 = monostable relay 2 c/o contacts
K1,K2,K3 = 6-way Mini-DIN socket (pins at 240°, PCB mount
S1,S2 = push-button (ITTD6-R)
Source: extremecircuits.net
When turning a computer on and off, various peripherals (such as printers, screen, scanner, etc.) often have to be turned on and off as well. By using the 5-V supply voltage from the USB interface on the PC, all these peripherals can easily be switched on and off at the same time as the PC. This principle can also be used with other appliances that have a USB interface (such as modern TVs and radios). This so-called ‘USB-standby-killer’ can be realized with just 5 components. The USB output voltage provides for the activation of the triac opto-driver (MOC3043) which has zero-crossing detection. This, in turn, drives the TRIAC, type BT126.

The circuit shown is used by the author for switching loads with a total power of about 150 W and is protected with a 1-A fuse. The circuit can easily handle much larger loads however. In that case and/or when using a very inductive load a so-called snubber network is required across the triac. The value of the fuse will also need to be changed as appropriate. The circuit can easily be built into a mains multi-way powerboard. Make sure you have good isolation between the USB and mains sections.

Circuit diagram:
USB Operated Home Appliances Circuit Diagram

Please don't make this circuit if you are not an expert!

USB Operated Home Appliances

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When turning a computer on and off, various peripherals (such as printers, screen, scanner, etc.) often have to be turned on and off as well. By using the 5-V supply voltage from the USB interface on the PC, all these peripherals can easily be switched on and off at the same time as the PC. This principle can also be used with other appliances that have a USB interface (such as modern TVs and radios). This so-called ‘USB-standby-killer’ can be realized with just 5 components. The USB output voltage provides for the activation of the triac opto-driver (MOC3043) which has zero-crossing detection. This, in turn, drives the TRIAC, type BT126.

The circuit shown is used by the author for switching loads with a total power of about 150 W and is protected with a 1-A fuse. The circuit can easily handle much larger loads however. In that case and/or when using a very inductive load a so-called snubber network is required across the triac. The value of the fuse will also need to be changed as appropriate. The circuit can easily be built into a mains multi-way powerboard. Make sure you have good isolation between the USB and mains sections.

Circuit diagram:
USB Operated Home Appliances Circuit Diagram

Please don't make this circuit if you are not an expert!
In normal suburban driving you pass through so many different speed zones that it can be a nuisance having to switch speed settings. The speed display can also be a distraction. This circuit eliminates the display and the need for speed selection. Each time you exceed a particular speed setting (eg, 40km/h, 50km/h, etc), a piezo buzzer will beep. Speed pulses are fed to the base of Q1 and the resulting waveform at its collector is fed via an RC network to the input of an LM2917 frequency-to-voltage converter, IC1. The resulting voltage is fed to three comparators (IC2d-IC2b) which have the reference voltages at their inverting inputs set by 10-turn trimpots VR1, VR2 & VR3. The output of each comparator is applied via another RC network to the gate of an SCR. The anodes of the three SCRs are commoned connected to the inverting input of the remaining comparator, IC2a.

Circuit diagram:
Speed Alarm Circuit Diagram

Its non-inverting input is set to +2.3V by trimpot VR4. In use, once you exceed the speed setting for a particular comparator, its associated SCR briefly conducts to pull pin 2 of IC2a low and a short beep is emitted by the piezo buzzer. Then, as you exceed the next speed setting, another beep will be heard. The idea is make each speed setting a few km/h higher than actual so that if you are driving at the correct speed in a given zone, the buzzer will not sound. But as you increase speed, the buzzer will beep once as you exceed the speed setting for each zone. In this way, there is no need to continually switch speed settings as you drive through different zones and you can choose to ignore beeps that are not "illegal".
Author: Col Edwards - Copyright: Silicon Chip Electronics Magazine

Speed Alarm For Cars

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In normal suburban driving you pass through so many different speed zones that it can be a nuisance having to switch speed settings. The speed display can also be a distraction. This circuit eliminates the display and the need for speed selection. Each time you exceed a particular speed setting (eg, 40km/h, 50km/h, etc), a piezo buzzer will beep. Speed pulses are fed to the base of Q1 and the resulting waveform at its collector is fed via an RC network to the input of an LM2917 frequency-to-voltage converter, IC1. The resulting voltage is fed to three comparators (IC2d-IC2b) which have the reference voltages at their inverting inputs set by 10-turn trimpots VR1, VR2 & VR3. The output of each comparator is applied via another RC network to the gate of an SCR. The anodes of the three SCRs are commoned connected to the inverting input of the remaining comparator, IC2a.

Circuit diagram:
Speed Alarm Circuit Diagram

Its non-inverting input is set to +2.3V by trimpot VR4. In use, once you exceed the speed setting for a particular comparator, its associated SCR briefly conducts to pull pin 2 of IC2a low and a short beep is emitted by the piezo buzzer. Then, as you exceed the next speed setting, another beep will be heard. The idea is make each speed setting a few km/h higher than actual so that if you are driving at the correct speed in a given zone, the buzzer will not sound. But as you increase speed, the buzzer will beep once as you exceed the speed setting for each zone. In this way, there is no need to continually switch speed settings as you drive through different zones and you can choose to ignore beeps that are not "illegal".
Author: Col Edwards - Copyright: Silicon Chip Electronics Magazine
The oscillator shown in Figure1 is frequently used in digital circuits and may, therefore, look very familiar. Many readers may not know that this type of oscillator suffers from a nasty draw-back caused by noise. When the amplitude of the noise is higher than the hysteresis of the gates used for the oscillator, spurious switching pulses are generated near the zero crossings. This problem can be cured only by ensuring that the rise time of the input signal is shorter than the reaction time of the relevant gate. When the oscillator is built with fast logic gates, such as those in the HC-series, the like-lihood of the problem occurring is great.


However, as long as the positive feedback is fast enough, nothing untoward will happen. However, when delays occur owing to the transit time of the components used, the problem may rear its head. In the configuration of Figure 1a, the signal passes through two inverters and thus experiences twice the transit time of a single gate. The upper signal in the oscilloscope trace in Figure 2 shows the result of this: the gates used are simply too fast for this type of oscillator. If one of the inverters is replaced by a buffer, and the oscillator is modified as shown in Figure 1b, the transit time is limited to that of one gate: the lower trace in Figure 2 shows that the oscillator then works correctly. The practical circuit diagram of the general-purpose oscillator is shown in Figure 3. Note that two XOR gates are used to ensure that the transit time of the buffer is equal to that of the inverter.

General Purpose Oscillator

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The oscillator shown in Figure1 is frequently used in digital circuits and may, therefore, look very familiar. Many readers may not know that this type of oscillator suffers from a nasty draw-back caused by noise. When the amplitude of the noise is higher than the hysteresis of the gates used for the oscillator, spurious switching pulses are generated near the zero crossings. This problem can be cured only by ensuring that the rise time of the input signal is shorter than the reaction time of the relevant gate. When the oscillator is built with fast logic gates, such as those in the HC-series, the like-lihood of the problem occurring is great.


However, as long as the positive feedback is fast enough, nothing untoward will happen. However, when delays occur owing to the transit time of the components used, the problem may rear its head. In the configuration of Figure 1a, the signal passes through two inverters and thus experiences twice the transit time of a single gate. The upper signal in the oscilloscope trace in Figure 2 shows the result of this: the gates used are simply too fast for this type of oscillator. If one of the inverters is replaced by a buffer, and the oscillator is modified as shown in Figure 1b, the transit time is limited to that of one gate: the lower trace in Figure 2 shows that the oscillator then works correctly. The practical circuit diagram of the general-purpose oscillator is shown in Figure 3. Note that two XOR gates are used to ensure that the transit time of the buffer is equal to that of the inverter.
Here is a Simple LED Volt meter to Monitor the charge level in Lead Acid Battery or Tubular battery. The terminal voltage of the battery is indicated through a four level LED indicators. The nominal terminal voltage of a Lead Acid battery is 13.8 volts and that of a Tubular battery is 14.8 volts when fully charged. The LED voltmeter uses four Zener diodes to light the LEDs at the precise breakdown voltage of the Zener diodes. Usually the Zener diode requires 1.6 volts in excess than its prescribed value to reach the breakdown threshold level. When the battery holds 13.6 volts or more, all the Zener breakdown and all LEDs light up. When the battery is discharged below 10.6 volts, all the LEDs remain dark. So depending on the terminal voltage of the battery, LEDs light up one by one or turns off.
LED Volt Meter Circuit

Circuit diagram:

LED Volt Meter Circuit Diagram
Author: D. Mohan Kumar
Copyright: electroschematics.com

LED Volt Meter

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Here is a Simple LED Volt meter to Monitor the charge level in Lead Acid Battery or Tubular battery. The terminal voltage of the battery is indicated through a four level LED indicators. The nominal terminal voltage of a Lead Acid battery is 13.8 volts and that of a Tubular battery is 14.8 volts when fully charged. The LED voltmeter uses four Zener diodes to light the LEDs at the precise breakdown voltage of the Zener diodes. Usually the Zener diode requires 1.6 volts in excess than its prescribed value to reach the breakdown threshold level. When the battery holds 13.6 volts or more, all the Zener breakdown and all LEDs light up. When the battery is discharged below 10.6 volts, all the LEDs remain dark. So depending on the terminal voltage of the battery, LEDs light up one by one or turns off.
LED Volt Meter Circuit

Circuit diagram:

LED Volt Meter Circuit Diagram
Author: D. Mohan Kumar
Copyright: electroschematics.com
Beeper or flashing LED alert, 1.5V battery powered portable unit

This circuit will emit an intermittent beep (or will flash a LED) when the water contained into a recipient has reached the desired level. It should be mounted on top of the recipient (e.g. a plastic tank) by means of two crocodile clips, acting also as probes. If a deeper sensing level is needed, the clips can be extended by means of two pieces of stiff wire (see pictures).

Circuit operation:
IC1, a 555 CMos timer chip, is wired as an astable multivibrator whose operating frequency is set by C1, R1 and R2, plus the resistance presented by water across the probes. If the resistance across the probes is zero (i.e. probes shorted), the output frequency will be about 3Hz and the sounder will beep (or the LED will flash) about three times per second. As water usually presents a certain amount of resistance, the actual oscillation frequency will be lower: less than one beep/flash per second. As probes will be increasingly immersed in water, the resistance across them will decrease and the oscillation frequency of IC1 will increase.

This means that a rough aural or visual indication of the level reached by water will be available. If a LED is chosen as the alert, C2, D1 and D2 must be added to the circuit in order to double the output voltage, thus allowing proper LED operation (see the rightmost part of the schematics). Interesting features of this circuit are 1.5V supply and ultra-low current consumption: 40µA in stand-by and 0.5mA in operation. This allows a single AAA alkaline cell to last several years and the saving of the power on/off switch.

Pictures of the project:

Screenshoot - Water Level Alert Circuit Schematic


Circuit diagram:
Water Level Alert Circuit Diagram

Parts:

R1 = 1K - 1/4W Resistor
R2 = 100K - 1/4W Resistor (See Notes)
C1 = 2.2uF-50V Electrolytic Capacitor
C2 = 220µF - 25V Electrolytic Capacitor (See Notes)
D1 = 5 or 10mm. Ultra-bright red LED (See Notes)
D2 = 1N5819 - 40V 1A Schottky-barrier Diode (See Notes)
IC = 7555 or TS555CN CMos Timer IC
BZ = Piezo sounder (incorporating 3KHz oscillator)
B1 = 1.5V Battery (AAA or AA cell etc.)
Two small crocodile clips
Two pieces of stiff wire of suitable length
Battery socket, etc.

Notes:
  • If a LED alert is needed instead of the beeper, R2 value must be changed to 10K, the Piezo sounder can be omitted and D1, D2 and C2 must be added, as shown in the rightmost part of the schematics.
  • A common red LED can be used for D1, but ultra-bright types are preferred.
  • Any Schottky-barrier type diode can be used in place of the 1N5819, e.g. the BAT46, rated @ 100V 150mA.
  • Wipe the probes regularly to avoid excessive resistance variations due to partial oxidization.

Water Level Alert Circuit Schematic

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Beeper or flashing LED alert, 1.5V battery powered portable unit

This circuit will emit an intermittent beep (or will flash a LED) when the water contained into a recipient has reached the desired level. It should be mounted on top of the recipient (e.g. a plastic tank) by means of two crocodile clips, acting also as probes. If a deeper sensing level is needed, the clips can be extended by means of two pieces of stiff wire (see pictures).

Circuit operation:
IC1, a 555 CMos timer chip, is wired as an astable multivibrator whose operating frequency is set by C1, R1 and R2, plus the resistance presented by water across the probes. If the resistance across the probes is zero (i.e. probes shorted), the output frequency will be about 3Hz and the sounder will beep (or the LED will flash) about three times per second. As water usually presents a certain amount of resistance, the actual oscillation frequency will be lower: less than one beep/flash per second. As probes will be increasingly immersed in water, the resistance across them will decrease and the oscillation frequency of IC1 will increase.

This means that a rough aural or visual indication of the level reached by water will be available. If a LED is chosen as the alert, C2, D1 and D2 must be added to the circuit in order to double the output voltage, thus allowing proper LED operation (see the rightmost part of the schematics). Interesting features of this circuit are 1.5V supply and ultra-low current consumption: 40µA in stand-by and 0.5mA in operation. This allows a single AAA alkaline cell to last several years and the saving of the power on/off switch.

Pictures of the project:

Screenshoot - Water Level Alert Circuit Schematic


Circuit diagram:
Water Level Alert Circuit Diagram

Parts:

R1 = 1K - 1/4W Resistor
R2 = 100K - 1/4W Resistor (See Notes)
C1 = 2.2uF-50V Electrolytic Capacitor
C2 = 220µF - 25V Electrolytic Capacitor (See Notes)
D1 = 5 or 10mm. Ultra-bright red LED (See Notes)
D2 = 1N5819 - 40V 1A Schottky-barrier Diode (See Notes)
IC = 7555 or TS555CN CMos Timer IC
BZ = Piezo sounder (incorporating 3KHz oscillator)
B1 = 1.5V Battery (AAA or AA cell etc.)
Two small crocodile clips
Two pieces of stiff wire of suitable length
Battery socket, etc.

Notes:
  • If a LED alert is needed instead of the beeper, R2 value must be changed to 10K, the Piezo sounder can be omitted and D1, D2 and C2 must be added, as shown in the rightmost part of the schematics.
  • A common red LED can be used for D1, but ultra-bright types are preferred.
  • Any Schottky-barrier type diode can be used in place of the 1N5819, e.g. the BAT46, rated @ 100V 150mA.
  • Wipe the probes regularly to avoid excessive resistance variations due to partial oxidization.
This circuit was designed to assist the installation of TV antennas. The signal is monitored using a small portable TV set and this circuit monitors the output of the TV's FM detector IC via a shielded lead. To initially calibrate the meter, adjust trimpot VR2 to zero the meter. Trimpot VR1 is a sensitivity control and can be set for a preset reading (ie, 0dB) or can be calibrated in millivolts. Rotating the antenna for a minimum reading on the meter (indicating FM quieting) gives the optimum orientation for the antenna.


Circuit diagram:

TV Relative Signal Strength Meter Circuit Diagram
Author: Ted Sherman
Copyright: Silicon Chip Electronics

TV Relative Signal Strength Meter

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This circuit was designed to assist the installation of TV antennas. The signal is monitored using a small portable TV set and this circuit monitors the output of the TV's FM detector IC via a shielded lead. To initially calibrate the meter, adjust trimpot VR2 to zero the meter. Trimpot VR1 is a sensitivity control and can be set for a preset reading (ie, 0dB) or can be calibrated in millivolts. Rotating the antenna for a minimum reading on the meter (indicating FM quieting) gives the optimum orientation for the antenna.


Circuit diagram:

TV Relative Signal Strength Meter Circuit Diagram
Author: Ted Sherman
Copyright: Silicon Chip Electronics
‘Hello… HELLO! Are you deaf? Do you have disco ears?’ If people ask you this and you’re still well below 80 , you may be suffering from hearing loss, which can come from (prolonged) listening to very loud music. You won’t notice how bad it is until it’s too late, and after that you won’t be able to hear your favorite music the way it really is – so an expensive sound system is no longer a sound investment. To avoid all this, use the i-trixx sound meter to save your ears (and your neighbor's ears!).

With just a handful of components, you can build a simple but effective sound level meter for your sound system. This sort of circuit is also called a VU meter. The abbreviation ‘VU’ stands for ‘volume unit’, which is used to express the average value of a music signal over a short time. The VU meter described here is what is called a ‘passive’ type. This means it does not need a separate power supply, since the power is provided by the input signal. This makes it easy to use: just connect it to the loudspeaker terminals (the polarity doesn’t matter) and you’re all set.

The more LEDs that light up while the music is playing, the more you should be asking yourself how well you are treating your ears (and your neighbours’ ears). Of course, this isn’t an accurately calibrated meter. The circuit design is too simple (and too inexpensive) for that. However, you can have a non-disco type (or your neighbors) tell you when the music is really too loud, and the maximum number of LED lit up at that time can serve you as a good reference for the maximum tolerable sound level.

Although this is a passive VU meter, it contains active components in the form of two transistors and six FETs. Seven LEDs light up in steps to show how much power is being pumped into the loudspeaker. The steps correspond to the power levels shown in the schematic for a sine-wave signal into an 8-ohm load. LED D1 lights up fi rst at low loudspeaker voltages. As the music power increases, the following LEDs (D2, D3, and so on) light up as well. The LEDs thus dance to the rhythm of the music (especially the bass notes).

Circuit diagram:

Noise Meter Circuit Diagram

This circuit can easily be assembled on a small piece of prototyping board. Use low-current types for the LEDs. They have a low forward voltage and are fairly bright at current levels as low as 1 mA. Connect the VU meter to the loudspeaker you want to monitor. If LED D2 never lights up (it remains dark even when LED D3 lights up), reverse the polarity of diode D8 (we have more to say about this later on). In addition, bear in mind that the sound from the speaker will have to be fairly loud before the LEDs will start lighting up.

If you want to know more about the technical details this VU meter, keep on reading. Each LED is driven by its own current source so it will not be overloaded with too much current when the input voltage increases. The current sources also ensure that the final amplifier is not loaded any more than necessary. The current sources for LEDs D1–D6 are formed by FET circuits. A FET can be made to supply a fixed current by simply connecting a resistor to the source lead (resistors R1–R6 in this case). With a resistance of 1 kΩ, the current is theoretically limited to 1 mA. However, in practice FETs have a especially broad tolerance range. The actual current level with our prototype ranged from 0.65 mA to 0.98 mA.

To ensure that each LED only lights up starting at a defined voltage, a Zener diode (D8–D13) is connected in series with each LED starting with D2. The Zener voltage must be approximately 3 V less than the voltage necessary for the indicated power level. The 3-V offset is a consequence of the voltage losses resulting from the LED, the FET, the rectifier, and the over voltage protection. The over voltage protection is combined with the current source for LED D7. One problem with using FETs as current sources is that the maximum rated drain–source voltage of the types used here is only 30 V.

If you want to use the circuit with an especially powerful fi nal amplifier, a maximum input level of slightly more than 30 V is much too low. We thus decided to double the limit. This job is handled by T7 and T8. If the amplitude of the applied signal is less than 30 V, T8 buffers the rectified voltage on C1. This means that when only the first LED is lit, the additional voltage drop of the over voltage protection circuit is primarily determined by the base–emitter voltage of T8. The maximum worst-case voltage drop across R8 is 0.7 V when all the LEDs are on, but it has increasingly less effect as the input voltage rises.

R8 is necessary so the base voltage can be regulated. R7 is fitted in series with LED D7 and Zener diode D13, and the voltage drop across R7 is used to cause transistor T7 to conduct. This voltage may be around 0.3 V at very low current levels, but with a current of a few mili-amperes it can be assumed to be 0.6 V. Transistor T7 starts conducting if the input voltage rises above the threshold voltage of D7 and D13, and this reduces the voltage on the base of T8. This negative feedback stabilizes the supply voltage for the LEDs at a level of around 30 V. With a value of 390 Ω for R7, the current through LED D7 will be slightly more than 1 mA.

This has been done intentionally so D7 will be a bit brighter than the other LEDs when the signal level is above 30 V. When the voltage is higher than 30 V, the circuit draws additional current due to the voltage drop across R8. The AC voltage on the loudspeaker terminals is half-wave rectifi ed by diode D14. This standard diode can handle 1 A at 400 V. The peak current level can be considerably higher, but don’t forget that the current still has to be provided by the fi nal amplifier.

Resistor R9 is included in series with the input to keep the additional load on the fi nal amplifi er within safe bounds and limit the interference or distortion that may result from this load. The peak current can never exceed 1.5 A (the charging current of C1), even when the circuit is connected directly to an AC voltage with an amplitude of 60 V. C1 also determines how long the LEDs stay lit. This brings us to an important aspect of the circuit, which you may wish to experiment with in combination with the current through the LEDs.

An important consideration in the circuit design is to keep the load on the fi nal amplifi er to a minimum. However, the combination of R9 and C1 causes an averaging of the complex music signal. The peak signal levels in the music are higher (or even much higher) than the average value. Tests made under actual conditions show that the applied peak power can easily be a factor of 2 to 4 greater than what is indicated by this VU meter. This amounts to 240 W or more with an 8-Ω loudspeaker.

You can reduce the value of C1 to make the circuit respond more quickly (and thus more accurately) to peak signal levels. Now a few comments on D8. You may receive a stabistor (for example, from the Philips BZV86 series or the like) for D8. Unlike a Zener diode, a stabistor must be connected in the forward-biased direction. A stabistor actually consists of a set of PN junctions in series (or ordinary forward-biased diodes). Check this carefully: if D2 does not light up when D8 is fi tted as a normal Zener diode, then D8 quite likely a stabistor, so you should fi t it the other way round.
Source: Elektor Electronics 12-2006

Save Your Ears - A Noise Meter Circuit

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‘Hello… HELLO! Are you deaf? Do you have disco ears?’ If people ask you this and you’re still well below 80 , you may be suffering from hearing loss, which can come from (prolonged) listening to very loud music. You won’t notice how bad it is until it’s too late, and after that you won’t be able to hear your favorite music the way it really is – so an expensive sound system is no longer a sound investment. To avoid all this, use the i-trixx sound meter to save your ears (and your neighbor's ears!).

With just a handful of components, you can build a simple but effective sound level meter for your sound system. This sort of circuit is also called a VU meter. The abbreviation ‘VU’ stands for ‘volume unit’, which is used to express the average value of a music signal over a short time. The VU meter described here is what is called a ‘passive’ type. This means it does not need a separate power supply, since the power is provided by the input signal. This makes it easy to use: just connect it to the loudspeaker terminals (the polarity doesn’t matter) and you’re all set.

The more LEDs that light up while the music is playing, the more you should be asking yourself how well you are treating your ears (and your neighbours’ ears). Of course, this isn’t an accurately calibrated meter. The circuit design is too simple (and too inexpensive) for that. However, you can have a non-disco type (or your neighbors) tell you when the music is really too loud, and the maximum number of LED lit up at that time can serve you as a good reference for the maximum tolerable sound level.

Although this is a passive VU meter, it contains active components in the form of two transistors and six FETs. Seven LEDs light up in steps to show how much power is being pumped into the loudspeaker. The steps correspond to the power levels shown in the schematic for a sine-wave signal into an 8-ohm load. LED D1 lights up fi rst at low loudspeaker voltages. As the music power increases, the following LEDs (D2, D3, and so on) light up as well. The LEDs thus dance to the rhythm of the music (especially the bass notes).

Circuit diagram:

Noise Meter Circuit Diagram

This circuit can easily be assembled on a small piece of prototyping board. Use low-current types for the LEDs. They have a low forward voltage and are fairly bright at current levels as low as 1 mA. Connect the VU meter to the loudspeaker you want to monitor. If LED D2 never lights up (it remains dark even when LED D3 lights up), reverse the polarity of diode D8 (we have more to say about this later on). In addition, bear in mind that the sound from the speaker will have to be fairly loud before the LEDs will start lighting up.

If you want to know more about the technical details this VU meter, keep on reading. Each LED is driven by its own current source so it will not be overloaded with too much current when the input voltage increases. The current sources also ensure that the final amplifier is not loaded any more than necessary. The current sources for LEDs D1–D6 are formed by FET circuits. A FET can be made to supply a fixed current by simply connecting a resistor to the source lead (resistors R1–R6 in this case). With a resistance of 1 kΩ, the current is theoretically limited to 1 mA. However, in practice FETs have a especially broad tolerance range. The actual current level with our prototype ranged from 0.65 mA to 0.98 mA.

To ensure that each LED only lights up starting at a defined voltage, a Zener diode (D8–D13) is connected in series with each LED starting with D2. The Zener voltage must be approximately 3 V less than the voltage necessary for the indicated power level. The 3-V offset is a consequence of the voltage losses resulting from the LED, the FET, the rectifier, and the over voltage protection. The over voltage protection is combined with the current source for LED D7. One problem with using FETs as current sources is that the maximum rated drain–source voltage of the types used here is only 30 V.

If you want to use the circuit with an especially powerful fi nal amplifier, a maximum input level of slightly more than 30 V is much too low. We thus decided to double the limit. This job is handled by T7 and T8. If the amplitude of the applied signal is less than 30 V, T8 buffers the rectified voltage on C1. This means that when only the first LED is lit, the additional voltage drop of the over voltage protection circuit is primarily determined by the base–emitter voltage of T8. The maximum worst-case voltage drop across R8 is 0.7 V when all the LEDs are on, but it has increasingly less effect as the input voltage rises.

R8 is necessary so the base voltage can be regulated. R7 is fitted in series with LED D7 and Zener diode D13, and the voltage drop across R7 is used to cause transistor T7 to conduct. This voltage may be around 0.3 V at very low current levels, but with a current of a few mili-amperes it can be assumed to be 0.6 V. Transistor T7 starts conducting if the input voltage rises above the threshold voltage of D7 and D13, and this reduces the voltage on the base of T8. This negative feedback stabilizes the supply voltage for the LEDs at a level of around 30 V. With a value of 390 Ω for R7, the current through LED D7 will be slightly more than 1 mA.

This has been done intentionally so D7 will be a bit brighter than the other LEDs when the signal level is above 30 V. When the voltage is higher than 30 V, the circuit draws additional current due to the voltage drop across R8. The AC voltage on the loudspeaker terminals is half-wave rectifi ed by diode D14. This standard diode can handle 1 A at 400 V. The peak current level can be considerably higher, but don’t forget that the current still has to be provided by the fi nal amplifier.

Resistor R9 is included in series with the input to keep the additional load on the fi nal amplifi er within safe bounds and limit the interference or distortion that may result from this load. The peak current can never exceed 1.5 A (the charging current of C1), even when the circuit is connected directly to an AC voltage with an amplitude of 60 V. C1 also determines how long the LEDs stay lit. This brings us to an important aspect of the circuit, which you may wish to experiment with in combination with the current through the LEDs.

An important consideration in the circuit design is to keep the load on the fi nal amplifi er to a minimum. However, the combination of R9 and C1 causes an averaging of the complex music signal. The peak signal levels in the music are higher (or even much higher) than the average value. Tests made under actual conditions show that the applied peak power can easily be a factor of 2 to 4 greater than what is indicated by this VU meter. This amounts to 240 W or more with an 8-Ω loudspeaker.

You can reduce the value of C1 to make the circuit respond more quickly (and thus more accurately) to peak signal levels. Now a few comments on D8. You may receive a stabistor (for example, from the Philips BZV86 series or the like) for D8. Unlike a Zener diode, a stabistor must be connected in the forward-biased direction. A stabistor actually consists of a set of PN junctions in series (or ordinary forward-biased diodes). Check this carefully: if D2 does not light up when D8 is fi tted as a normal Zener diode, then D8 quite likely a stabistor, so you should fi t it the other way round.
Source: Elektor Electronics 12-2006
The alarm may be used for a variety of applications, such as frost monitor, room temperature monitor, and so on. In the quiescent state, the circuit draws a current of only a few microamperes, so that, in theory at least, a 9 V dry battery (PP3, 6AM6, MN1604, 6LR61) should last for up to ten years. Such a tiny current is not possible when ICs are used, and the circuit is therefore a discrete design. Every four seconds a measuring bridge, which actuates a Schmitt trigger, is switched on for 150 ms by a clock generator. In that period of 150 ms, the resistance of an NTC thermistor, R11, is compared with that of a fixed resistor. If the former is less than the latter, the alarm is set off.

When the circuit is switched on, capacitor C1 is not charged and transistors T1–T3 are off. After switch-on, C1 is charged gradually via R1, R7, and R8, until the base voltage of T1 exceeds the threshold bias. Transistor T1 then comes on and causes T2 and T3 to conduct also. Thereupon, C1 is charged via current source T1-T2-D1, until the current from the source becomes smaller than that flowing through R3 and T3 (about 3 µA). This results in T1 switching off, so that, owing to the coupling with C1, the entire circuit is disabled. Capacitor C1 is (almost) fully charged, so that the anode potential of D1 drops well below 0 V. Only when C1 is charged again can a new cycle begin.

Circuit Diagram:

General-Purpose Alarm Circuit Diagram

It is obvious that the larger part of the current is used for charging C1. Gate IC1a functions as impedance inverter and feedback stage, and regularly switches on measurement bridge R9–R12-C2-P1 briefly. The bridge is terminated in a differential amplifier, which, in spite of the tiny current (and the consequent small transconductance of the transistors) provides a large amplification and, therefore, a high sensitivity. Resistors R13 and R15 provide through a kind of hysteresis a Schmitt trigger input for the differential amplifier, which results in unambiguous and fast measurement results. Capacitor C2 compensates for the capacitive effect of long cables between sensor and circuit and so prevents false alarms.

If the sensor (R11) is built in the same enclosure as the remainder of the circuit (as, for instance, in a room temperature monitor), C2 and R13 may be omitted. In that case,C3 willabsorb any interference signals and so prevent false alarms. To prevent any residual charge in C3 causing a false alarm when the bridge is in equilibrium, the capacitor is discharged rapidly via D2 when this happens. Gates IC1c and IC1d form an oscillator to drive the buzzer (an a.c. type). Owing to the very high impedance of the clock, an epoxy resin (not pertinax) board must be used for building the alarm. For the same reason, C1 should be a type with very low leakage current. If operation of the alarm is required when the resistance of R11 is higher than that of the fixed resistor, reverse the connections of the elements of the bridge and thus effectively the inverting and non-inverting inputs of the differential amplifier.

An NTC thermistor such as R11 has a resistance at –18 °C that is about ten times as high as that at room temperature. It is, therefore, advisable, if not a must, when precise operation is required, to consult the data sheet of the device or take a number of test readings. For the present circuit, the resistance at –18 °C must be 300–400 kΩ. The value of R12 should be the same. Preset P1 provides fine adjustment of the response threshold. Note that although the prototype uses an NTC thermistor, a different kind of sensor may also be used, provided its electrical specification is known and suits the present circuit.
Author: K. Syttkus
Copyright: Elektor Electronics

General-Purpose Alarm

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The alarm may be used for a variety of applications, such as frost monitor, room temperature monitor, and so on. In the quiescent state, the circuit draws a current of only a few microamperes, so that, in theory at least, a 9 V dry battery (PP3, 6AM6, MN1604, 6LR61) should last for up to ten years. Such a tiny current is not possible when ICs are used, and the circuit is therefore a discrete design. Every four seconds a measuring bridge, which actuates a Schmitt trigger, is switched on for 150 ms by a clock generator. In that period of 150 ms, the resistance of an NTC thermistor, R11, is compared with that of a fixed resistor. If the former is less than the latter, the alarm is set off.

When the circuit is switched on, capacitor C1 is not charged and transistors T1–T3 are off. After switch-on, C1 is charged gradually via R1, R7, and R8, until the base voltage of T1 exceeds the threshold bias. Transistor T1 then comes on and causes T2 and T3 to conduct also. Thereupon, C1 is charged via current source T1-T2-D1, until the current from the source becomes smaller than that flowing through R3 and T3 (about 3 µA). This results in T1 switching off, so that, owing to the coupling with C1, the entire circuit is disabled. Capacitor C1 is (almost) fully charged, so that the anode potential of D1 drops well below 0 V. Only when C1 is charged again can a new cycle begin.

Circuit Diagram:

General-Purpose Alarm Circuit Diagram

It is obvious that the larger part of the current is used for charging C1. Gate IC1a functions as impedance inverter and feedback stage, and regularly switches on measurement bridge R9–R12-C2-P1 briefly. The bridge is terminated in a differential amplifier, which, in spite of the tiny current (and the consequent small transconductance of the transistors) provides a large amplification and, therefore, a high sensitivity. Resistors R13 and R15 provide through a kind of hysteresis a Schmitt trigger input for the differential amplifier, which results in unambiguous and fast measurement results. Capacitor C2 compensates for the capacitive effect of long cables between sensor and circuit and so prevents false alarms.

If the sensor (R11) is built in the same enclosure as the remainder of the circuit (as, for instance, in a room temperature monitor), C2 and R13 may be omitted. In that case,C3 willabsorb any interference signals and so prevent false alarms. To prevent any residual charge in C3 causing a false alarm when the bridge is in equilibrium, the capacitor is discharged rapidly via D2 when this happens. Gates IC1c and IC1d form an oscillator to drive the buzzer (an a.c. type). Owing to the very high impedance of the clock, an epoxy resin (not pertinax) board must be used for building the alarm. For the same reason, C1 should be a type with very low leakage current. If operation of the alarm is required when the resistance of R11 is higher than that of the fixed resistor, reverse the connections of the elements of the bridge and thus effectively the inverting and non-inverting inputs of the differential amplifier.

An NTC thermistor such as R11 has a resistance at –18 °C that is about ten times as high as that at room temperature. It is, therefore, advisable, if not a must, when precise operation is required, to consult the data sheet of the device or take a number of test readings. For the present circuit, the resistance at –18 °C must be 300–400 kΩ. The value of R12 should be the same. Preset P1 provides fine adjustment of the response threshold. Note that although the prototype uses an NTC thermistor, a different kind of sensor may also be used, provided its electrical specification is known and suits the present circuit.
Author: K. Syttkus
Copyright: Elektor Electronics
Protect your valuable laptop against theft using this miniature alarm generator. Fixed in-side the laptop case, it will sound a loud alarm when someone tries to take the laptop. This highly sensitive circuit uses a homemade tilt switch to activate the alarm through tilting of the laptop case. The circuit uses readily available components and can be assembled on a small piece of Vero board or a general-purpose PCB.

It is powered by a 12V miniature battery used in remote control devices. IC TLO71 (IC1) is used as a voltage comparator with a potential divider comprising R2 and R3 providing half supply voltage at the non-inverting input (pin 3) of IC1. The inverting input receives a higher voltage through a water-activated tilt switch only when the probes in the tilt switch make contact with water.

When the tilt switch is kept in the horizontal position, the inverting input of IC1 gets a higher voltage than its non-inverting input and the output remains low. IC CD4538 (IC2) is used as a monostable with timing elements R5 and C1. With the shown values, the output of IC2 remains low for a period of three minutes. CD4538 is a precision monostable multivibrator free from false triggering and is more reliable than the popular timer IC 555.

Circuit diagram:
Laptop Protector Circuit Diagram

Its output becomes high when power is switched on and it becomes low when the trigger input (pin 5) gets a low-to-high transition pulse. The unit is fixed inside the laptop case in horizontal position. In this position, water inside the tilt switch effectively shorts the contacts, so the output of IC1 remains low. The alarm generator remains silent in the standby mode as trigger pin 5 of IC2 is low.

When someone tries to take the laptop case, the unit takes the vertical position and the tilt switch breaks the electrical contact between the probes Immediately the output of IC1 becomes high and monostable IC2 is triggered. The low output from IC2 triggers the pnp transistor (T1) and the buzzer starts beeping. Assemble the circuit as compactly as possible so as to make the unit matchbox size.

Make the tilt switch using a small (2.5cm long and 1cm wide) plastic bottle with two stainless pins as contacts. Fill two-third of the bottle with water such that the contacts never make electrical path when the tilt switch is in vertical position. Make the bottle leak-proof with adhesive or wax. Fix the tilt switch inside the enclosure of the circuit in horizontal position.

Fit the unit inside the laptop case in horizontal position using adhesive. Use a miniature buzzer and a micro switch (S1) to make the gadget compact. Keep the laptop case in horizontal position and switch on the unit. Your laptop is now protected.
Source: EFY Mag

Laptop Protector

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Protect your valuable laptop against theft using this miniature alarm generator. Fixed in-side the laptop case, it will sound a loud alarm when someone tries to take the laptop. This highly sensitive circuit uses a homemade tilt switch to activate the alarm through tilting of the laptop case. The circuit uses readily available components and can be assembled on a small piece of Vero board or a general-purpose PCB.

It is powered by a 12V miniature battery used in remote control devices. IC TLO71 (IC1) is used as a voltage comparator with a potential divider comprising R2 and R3 providing half supply voltage at the non-inverting input (pin 3) of IC1. The inverting input receives a higher voltage through a water-activated tilt switch only when the probes in the tilt switch make contact with water.

When the tilt switch is kept in the horizontal position, the inverting input of IC1 gets a higher voltage than its non-inverting input and the output remains low. IC CD4538 (IC2) is used as a monostable with timing elements R5 and C1. With the shown values, the output of IC2 remains low for a period of three minutes. CD4538 is a precision monostable multivibrator free from false triggering and is more reliable than the popular timer IC 555.

Circuit diagram:
Laptop Protector Circuit Diagram

Its output becomes high when power is switched on and it becomes low when the trigger input (pin 5) gets a low-to-high transition pulse. The unit is fixed inside the laptop case in horizontal position. In this position, water inside the tilt switch effectively shorts the contacts, so the output of IC1 remains low. The alarm generator remains silent in the standby mode as trigger pin 5 of IC2 is low.

When someone tries to take the laptop case, the unit takes the vertical position and the tilt switch breaks the electrical contact between the probes Immediately the output of IC1 becomes high and monostable IC2 is triggered. The low output from IC2 triggers the pnp transistor (T1) and the buzzer starts beeping. Assemble the circuit as compactly as possible so as to make the unit matchbox size.

Make the tilt switch using a small (2.5cm long and 1cm wide) plastic bottle with two stainless pins as contacts. Fill two-third of the bottle with water such that the contacts never make electrical path when the tilt switch is in vertical position. Make the bottle leak-proof with adhesive or wax. Fix the tilt switch inside the enclosure of the circuit in horizontal position.

Fit the unit inside the laptop case in horizontal position using adhesive. Use a miniature buzzer and a micro switch (S1) to make the gadget compact. Keep the laptop case in horizontal position and switch on the unit. Your laptop is now protected.
Source: EFY Mag
Although you may well be the proud owner of the very latest NiCd battery charger, you may still come across the odd 'incompatible' battery, for example, one having a rare voltage or requiring a much higher charging current than can be supplied by your off-the-shelf charger. In these cases, many of you will resort to an adjustable mains adaptor (say, a 500-mA type) because that is probably the cheapest way of providing the direct voltage required to charge the battery. Not fast and not very efficient, this 'rustic' charging system works, although subject to the following restrictions:

Circuit diagram:

Battery-Charging Indicator Circuit Diagram
  1. You should have some idea of the charging current. In case you use an adaptor which is adjustable but of the unregulated, low output current type, you can adjust the current by adjusting the output voltage.
  2. You have to know if the current actually flows through the battery. A current-detecting indicator is therefore much to be preferred over a voltage indicator.
  3. To prevent you from forgetting all about the charging cycle, the indicator should be visible from wherever you pass by frequently. Using the circuit shown here, the LED lights when the baseemitter potential of the transistor exceeds about 0.2 V. Using a resistor of 1 ? as suggested this happens at a current of about 200 mA, or about 40 mA if R1 is changed to 4.7?. The voltage drop caused by this indicator can never exceed the base-emitter voltage (UBE) of the transistor, or about 0.7V. Even if the current through R1 continues to increase beyond the level at which UBE = 0.7 V, the base of the transistor will 'absorb' the excess current. The TO-220 style BU406 transistor suggested here is capable of accepting base currents up to 4A. Using this charging indicator you have overcome the restrictions 2 and 3 mentioned above.

Battery-Charging Indicator For Mains Adaptor

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Although you may well be the proud owner of the very latest NiCd battery charger, you may still come across the odd 'incompatible' battery, for example, one having a rare voltage or requiring a much higher charging current than can be supplied by your off-the-shelf charger. In these cases, many of you will resort to an adjustable mains adaptor (say, a 500-mA type) because that is probably the cheapest way of providing the direct voltage required to charge the battery. Not fast and not very efficient, this 'rustic' charging system works, although subject to the following restrictions:

Circuit diagram:

Battery-Charging Indicator Circuit Diagram
  1. You should have some idea of the charging current. In case you use an adaptor which is adjustable but of the unregulated, low output current type, you can adjust the current by adjusting the output voltage.
  2. You have to know if the current actually flows through the battery. A current-detecting indicator is therefore much to be preferred over a voltage indicator.
  3. To prevent you from forgetting all about the charging cycle, the indicator should be visible from wherever you pass by frequently. Using the circuit shown here, the LED lights when the baseemitter potential of the transistor exceeds about 0.2 V. Using a resistor of 1 ? as suggested this happens at a current of about 200 mA, or about 40 mA if R1 is changed to 4.7?. The voltage drop caused by this indicator can never exceed the base-emitter voltage (UBE) of the transistor, or about 0.7V. Even if the current through R1 continues to increase beyond the level at which UBE = 0.7 V, the base of the transistor will 'absorb' the excess current. The TO-220 style BU406 transistor suggested here is capable of accepting base currents up to 4A. Using this charging indicator you have overcome the restrictions 2 and 3 mentioned above.
My new home theatre receiver was getting rather hot in the close confines of its cabinet, with the temperature reaching over 40°C after only about 30 minutes of use. To help lower the temperature, I decided to install a fan in the cabinet. A 75mm hole was cut in the shelf under the receiver, and a 12V fan salvaged from an old computer power supply was mounted underneath. The fan was powered from a 12V DC plugpack. This did the job, keeping the temperature below 30°C even after prolonged use on a warm day. However, the fan was annoyingly loud when running at full speed. To reduce the noise level substantially, I built this fan speed controller with temperature feedback.

The circuit was culled from variety of ideas found on various sites on the internet, with the final circuit designed from what was in the "junk box". Air temperature in the cabinet is sensed via an LM335 (TS1). It is glued to a piece of aluminium about 25mm square with instant glue, which is then attached to the top of the receiver with "Blue-Tack". About 300mm of audio coax makes the connection back to the circuit board. The LM335’s output rises 10mV per degree Centigrade. It is calibrated to zero output at -273°C, so at 20°C, the output will be 2.93V. This is applied to the non-inverting input of a 741 op amp (IC1).

Circuit diagram:
Junk-box Fan Speed Controller Circuit Diagram

A 1N4733 5.1V Zener diode provides a voltage reference for the inverting input via trimpot VR1. The output of the op amp drives a TIP122 Darlington transistor (Q1), which in turn drives the fan motor. The op amp gain was calculated to give about 12V to the fan at 40°C. To keep the transistor cool, it is mounted on the metal base of a small plastic box, which is also used to house the components. Initial setup should be performed with everything turned off and the ambient temperature at about 20°C. Adjust the 10-turn pot until the fan just stops running. I used a gasket made from foam strips and "blue-tacked" them between the feet of the receiver to direct all of the airflow through it. The temperature now remains at about 32°C, the fan runs very quietly and continues to run down for about 30 minutes after the receiver is switched off.
Author: Martin Cook - Copyright: Silicon Chip Electronics

Junk-box Fan Speed Controller

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My new home theatre receiver was getting rather hot in the close confines of its cabinet, with the temperature reaching over 40°C after only about 30 minutes of use. To help lower the temperature, I decided to install a fan in the cabinet. A 75mm hole was cut in the shelf under the receiver, and a 12V fan salvaged from an old computer power supply was mounted underneath. The fan was powered from a 12V DC plugpack. This did the job, keeping the temperature below 30°C even after prolonged use on a warm day. However, the fan was annoyingly loud when running at full speed. To reduce the noise level substantially, I built this fan speed controller with temperature feedback.

The circuit was culled from variety of ideas found on various sites on the internet, with the final circuit designed from what was in the "junk box". Air temperature in the cabinet is sensed via an LM335 (TS1). It is glued to a piece of aluminium about 25mm square with instant glue, which is then attached to the top of the receiver with "Blue-Tack". About 300mm of audio coax makes the connection back to the circuit board. The LM335’s output rises 10mV per degree Centigrade. It is calibrated to zero output at -273°C, so at 20°C, the output will be 2.93V. This is applied to the non-inverting input of a 741 op amp (IC1).

Circuit diagram:
Junk-box Fan Speed Controller Circuit Diagram

A 1N4733 5.1V Zener diode provides a voltage reference for the inverting input via trimpot VR1. The output of the op amp drives a TIP122 Darlington transistor (Q1), which in turn drives the fan motor. The op amp gain was calculated to give about 12V to the fan at 40°C. To keep the transistor cool, it is mounted on the metal base of a small plastic box, which is also used to house the components. Initial setup should be performed with everything turned off and the ambient temperature at about 20°C. Adjust the 10-turn pot until the fan just stops running. I used a gasket made from foam strips and "blue-tacked" them between the feet of the receiver to direct all of the airflow through it. The temperature now remains at about 32°C, the fan runs very quietly and continues to run down for about 30 minutes after the receiver is switched off.
Author: Martin Cook - Copyright: Silicon Chip Electronics
This circuit uses a step-up switch-mode regulator, which is usually used to produce a positive supply, to generate a regulated negative output voltage. The device used here is the MIC4680 from Micrel (www.micrel.com), but the idea would of course work with similar regulators from other manufacturers. Because of coil L1, which performs the voltage conversion by the intermediate storage of energy in the form of a magnetic field, the output is effectively isolated from the input. We can therefore connect the right-hand side of L1 to ground rather than to the positive output without causing a large current to flow.



Then we connect the ground pin of the regulator IC and all the components connected to it as the negative voltage output, isolated from ground. The components on the output side of the regulator are connected as usual: flywheel diode D1, coil L1 and the voltage divider formed by R1 and R2. These last two components set the output voltage, according to a formula given in the data sheet. Example component values for the MIC4680 used here are given in the table. The input voltage should lie within the permitted range for the regulator used, and must in any case be at least as great in magnitude as the desired output voltage (here +5 V or +12 V), so that the step-down regulation technique can work.



It is important to take care when building this circuit to mount the regulator using an insulator, since generally the GND pin of the device is connected to the heatsink tab. Also, the ON/OFF control input cannot be driven using a normal logic signal, since the regulator’s ground reference is the output voltage rather than ground itself. If the ON/OFF function is required, a level shifter or optocoupler must be used.

Voltage Inverter Using Switch-Mode Regulator

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This circuit uses a step-up switch-mode regulator, which is usually used to produce a positive supply, to generate a regulated negative output voltage. The device used here is the MIC4680 from Micrel (www.micrel.com), but the idea would of course work with similar regulators from other manufacturers. Because of coil L1, which performs the voltage conversion by the intermediate storage of energy in the form of a magnetic field, the output is effectively isolated from the input. We can therefore connect the right-hand side of L1 to ground rather than to the positive output without causing a large current to flow.



Then we connect the ground pin of the regulator IC and all the components connected to it as the negative voltage output, isolated from ground. The components on the output side of the regulator are connected as usual: flywheel diode D1, coil L1 and the voltage divider formed by R1 and R2. These last two components set the output voltage, according to a formula given in the data sheet. Example component values for the MIC4680 used here are given in the table. The input voltage should lie within the permitted range for the regulator used, and must in any case be at least as great in magnitude as the desired output voltage (here +5 V or +12 V), so that the step-down regulation technique can work.



It is important to take care when building this circuit to mount the regulator using an insulator, since generally the GND pin of the device is connected to the heatsink tab. Also, the ON/OFF control input cannot be driven using a normal logic signal, since the regulator’s ground reference is the output voltage rather than ground itself. If the ON/OFF function is required, a level shifter or optocoupler must be used.
Charge Your Mobile Phone While Enjoying The Journey

Here is an ideal Mobile charger using 1.5 volt pen cells to charge mobile phone while traveling. It can replenish cell phone battery three or four times in places where AC power is not available. Most of the Mobile phone batteries are rated at 3.6 V/500 mA. A single pen torch cell can provide 1.5 volts and 1.5 Amps current. So if four pen cells are connected serially, it will form a battery pack with 6 volt and 1.5 Amps current. When power is applied to the circuit through S1, transistor Q1 conducts and Green LED lights.

When Q1 conducts Q2 also conducts since its base becomes negative. Charging current flows from the collector of Q1. To reduce the charging voltage to 4.7 volts, Zener diode D2 is used. The output gives 20 mA current for slow charging. If more current is required for fast charging, reduce the value of R4 to 47 ohms so that 80 mA current will be available. Output points are used to connect the charger with the mobile phone. Use suitable pins for this and connect with correct polarity. The circuit comes from here.

Circuit diagram:
Mobile Phone Travel Charger
Parts:

R1 = 1K
R2 = 470R
R3 = 4.7K
R4 = 270R
R5 = 27R
C1 = 100uF-25V
D1 = Green LED
D2 = 4.7V/1W Zener
B1 = 1.5Vx4 Cells
S1 = On/Off Switch
Q1 = BC548
Q2 = SK100

Mobile Phone Travel Charger Circuit Diagram

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Charge Your Mobile Phone While Enjoying The Journey

Here is an ideal Mobile charger using 1.5 volt pen cells to charge mobile phone while traveling. It can replenish cell phone battery three or four times in places where AC power is not available. Most of the Mobile phone batteries are rated at 3.6 V/500 mA. A single pen torch cell can provide 1.5 volts and 1.5 Amps current. So if four pen cells are connected serially, it will form a battery pack with 6 volt and 1.5 Amps current. When power is applied to the circuit through S1, transistor Q1 conducts and Green LED lights.

When Q1 conducts Q2 also conducts since its base becomes negative. Charging current flows from the collector of Q1. To reduce the charging voltage to 4.7 volts, Zener diode D2 is used. The output gives 20 mA current for slow charging. If more current is required for fast charging, reduce the value of R4 to 47 ohms so that 80 mA current will be available. Output points are used to connect the charger with the mobile phone. Use suitable pins for this and connect with correct polarity. The circuit comes from here.

Circuit diagram:
Mobile Phone Travel Charger
Parts:

R1 = 1K
R2 = 470R
R3 = 4.7K
R4 = 270R
R5 = 27R
C1 = 100uF-25V
D1 = Green LED
D2 = 4.7V/1W Zener
B1 = 1.5Vx4 Cells
S1 = On/Off Switch
Q1 = BC548
Q2 = SK100
There are monitors which only have three BNC inputs and which use composite synchronization (‘sync on green’). This circuit has been designed with these types of monitor in mind. As can be seen, the circuit has been kept very simple, but it still gives a reasonable performance. The principle of operation is very straightforward. The RGB signals from the VGA connector are fed to three BNC connectors via AC-coupling capacitors. These have been added to stop any direct current from entering the VGA card. A pull-up resistor on the green output provides a DC offset, while a transistor (a BS170 MOSFET) can switch this output to ground. It is possible to get synchronisation problems when the display is extremely bright, with a maximum green component.
In this case the value of R2 should be reduced a little, but this has the side effect that the brightness noticeably decreases and the load on the graphics card increases. To keep the colour balance the same, the resistors for the other two colors (R1 en R3) have to be changed to the same value as R2. An EXOR gate from IC1 (74HC86) combines the separate V-sync and H-sync signals into a composite sync signal. Since the sync in DOS-modes is often inverted compared to the modes commonly used by Windows, the output of IC1a is inverted by IC1b. JP1 can then by used to select the correct operating mode. This jumper can be replaced by a small two-way switch, if required.
This switch should be mounted directly onto the PCB, as any connecting wires will cause a lot of interference. The PCB has been kept as compact as possible, so the circuit can be mounted in a small metal (earthed!) enclosure. With a monitor connected the current consumption will be in the region of 30 mA. A 78L05 voltage regulator provides a stable 5 V, making it possible to use any type of mains adapter, as long as it supplies at least 9 V. Diode D2 provides protection against a reverse polarity. LED D1 indicates when the supply is present. The circuit should be powered up before connecting it to an active VGA output, as otherwise the sync signals will feed the circuit via the internal protection diodes of IC1, which can be noticed by a dimly lit LED. This is something best avoided.

Resistors:
R1,R2,R3 = 470Ω
R4 = 100Ω
R5 = 3kΩ3
Capacitors:
C1,C3,C5 = 47µF 25V radial
C2,C4,C6,C7,C10 = 100nF ceramic
C8 = 4µF7 63V radial
C9 = 100µF 25V radial
Semiconductors:
D1 = LED, high-efficiency
D2 = 1N4002
T1 = BS170
IC1 = 74HC86
IC2 = 78L05
Miscellaneous:
JP1 = 3-way pinheader with jumper
K1 = 15-way VGA socket (female), PCB mount (angled pins)
K2,K3,K4 = BNC socket (female), PCB mount, 75Ω

VGA to BNC Adapter (Converter)

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There are monitors which only have three BNC inputs and which use composite synchronization (‘sync on green’). This circuit has been designed with these types of monitor in mind. As can be seen, the circuit has been kept very simple, but it still gives a reasonable performance. The principle of operation is very straightforward. The RGB signals from the VGA connector are fed to three BNC connectors via AC-coupling capacitors. These have been added to stop any direct current from entering the VGA card. A pull-up resistor on the green output provides a DC offset, while a transistor (a BS170 MOSFET) can switch this output to ground. It is possible to get synchronisation problems when the display is extremely bright, with a maximum green component.
In this case the value of R2 should be reduced a little, but this has the side effect that the brightness noticeably decreases and the load on the graphics card increases. To keep the colour balance the same, the resistors for the other two colors (R1 en R3) have to be changed to the same value as R2. An EXOR gate from IC1 (74HC86) combines the separate V-sync and H-sync signals into a composite sync signal. Since the sync in DOS-modes is often inverted compared to the modes commonly used by Windows, the output of IC1a is inverted by IC1b. JP1 can then by used to select the correct operating mode. This jumper can be replaced by a small two-way switch, if required.
This switch should be mounted directly onto the PCB, as any connecting wires will cause a lot of interference. The PCB has been kept as compact as possible, so the circuit can be mounted in a small metal (earthed!) enclosure. With a monitor connected the current consumption will be in the region of 30 mA. A 78L05 voltage regulator provides a stable 5 V, making it possible to use any type of mains adapter, as long as it supplies at least 9 V. Diode D2 provides protection against a reverse polarity. LED D1 indicates when the supply is present. The circuit should be powered up before connecting it to an active VGA output, as otherwise the sync signals will feed the circuit via the internal protection diodes of IC1, which can be noticed by a dimly lit LED. This is something best avoided.

Resistors:
R1,R2,R3 = 470Ω
R4 = 100Ω
R5 = 3kΩ3
Capacitors:
C1,C3,C5 = 47µF 25V radial
C2,C4,C6,C7,C10 = 100nF ceramic
C8 = 4µF7 63V radial
C9 = 100µF 25V radial
Semiconductors:
D1 = LED, high-efficiency
D2 = 1N4002
T1 = BS170
IC1 = 74HC86
IC2 = 78L05
Miscellaneous:
JP1 = 3-way pinheader with jumper
K1 = 15-way VGA socket (female), PCB mount (angled pins)
K2,K3,K4 = BNC socket (female), PCB mount, 75Ω
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How often on average do you have to call members of your family each day to tell them that dinner is ready, it’s time to leave, and the like? The person you want is usually in a different room, such as the hobby room or bedroom. A powerful buzzer in the room, combined with a pushbutton at the bottom of the stairs or in the kitchen, could be very handy in such situations. The heart of this circuit is formed by IC1, a TDA2030. This IC has built-in thermal protection, so it’s not likely to quickly give up the ghost. R1 and R2 apply a voltage equal to half the supply voltage to the plus input of the opamp. R3 provides positive feedback. Finally, the combination of C2, R4 and trimmer P12 determines the oscillation frequency of the circuit.


 Power Buzzer Circuit Diagram


The frequency of the tone can also be adjusted using P1. There is no volume control, since you always want to get attention when you press pushbutton S1. Fit the entire circuit where you want to have the pushbutton. The loudspeaker can then be placed in a strategic location, such as in the bedroom or wherever is appropriate. Use speaker cable to connect the loudspeaker. Normal bell wire can cause a significant power loss if the loudspeaker is relatively far away. The loudspeaker must be able to handle a continuous power of at least 6 W (with a 20-V supply voltage).

The power quickly drops as the supply voltage decreases (P = Urms 2 / RL). The power supply for this circuit is not particularly critical. However, it must be able to provide sufficient current. A good nominal value is around 400 mA at 20 V. At 4 V, it will be approximately 25 mA. Most likely, you can find a suitable power supply somewhere in your hobby room. Otherwise, you can certainly find a low-cost power supply design in our circuits archive that will fill the bill!
Author: G. Baars
Copyright: Elektor Electronics

Power Buzzer

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How often on average do you have to call members of your family each day to tell them that dinner is ready, it’s time to leave, and the like? The person you want is usually in a different room, such as the hobby room or bedroom. A powerful buzzer in the room, combined with a pushbutton at the bottom of the stairs or in the kitchen, could be very handy in such situations. The heart of this circuit is formed by IC1, a TDA2030. This IC has built-in thermal protection, so it’s not likely to quickly give up the ghost. R1 and R2 apply a voltage equal to half the supply voltage to the plus input of the opamp. R3 provides positive feedback. Finally, the combination of C2, R4 and trimmer P12 determines the oscillation frequency of the circuit.


 Power Buzzer Circuit Diagram


The frequency of the tone can also be adjusted using P1. There is no volume control, since you always want to get attention when you press pushbutton S1. Fit the entire circuit where you want to have the pushbutton. The loudspeaker can then be placed in a strategic location, such as in the bedroom or wherever is appropriate. Use speaker cable to connect the loudspeaker. Normal bell wire can cause a significant power loss if the loudspeaker is relatively far away. The loudspeaker must be able to handle a continuous power of at least 6 W (with a 20-V supply voltage).

The power quickly drops as the supply voltage decreases (P = Urms 2 / RL). The power supply for this circuit is not particularly critical. However, it must be able to provide sufficient current. A good nominal value is around 400 mA at 20 V. At 4 V, it will be approximately 25 mA. Most likely, you can find a suitable power supply somewhere in your hobby room. Otherwise, you can certainly find a low-cost power supply design in our circuits archive that will fill the bill!
Author: G. Baars
Copyright: Elektor Electronics
Although it does not have the same charm as real mercury barometers with long glass tubes on pieces of carved and polished wood, the Torricelli barometer discussed here is a functional equivalent and electronic replica of the Torricelli barometer. Actually, rather than displaying the atmospheric pressure on the traditional digital displays, we preferred to reproduce the general look of this respected predecessor of electronic barometers.

The mercury tube is, of course, replaced by a simple LED scale which, if not as beautiful, is still less toxic for the environment in case of breakage. As indicated on the drawing, the pressure sensor utilized is a Motorola MPX2200AP. This circuit is adapted for measuring absolute pressure and has a range well suited for atmospheric pressure. Without entering too deep into the technical details, such sensors deliver an output of voltage proportional not only to the measured pressure but, unfortunately, to their supply voltage as well.

Hence they must be powered from a stable voltage which is ensured here by the use of IC1. Since the output of the MPX2200 is differential and at a very low level, we had to resort to the use of four operational amplifiers IC4.A to IC4.D, contained in one LM324, to obtain levels that can be processed easily. As long as potentiometer P1 is adjusted correctly, this group of operational amplifiers delivers a voltage of 1 volt per atmospheric pressure of 1,000 hPa to the LM3914.


Circuit diagram:
Electronic Torricelli Barometer Circuit Diagram

Since the atmospheric pressure will be within the range 950 to 1040 hPa at sea level, we need to make an expanded-scale voltmeter with this LM3914 in order to better exploit the 10 LEDs that it can control. That is the role of resistors R7 and R8 which artificially raise the minimum voltage value the chip is capable of measuring. Consequently, we can ‘calibrate’ our LED scale with one LED per 10 hPa and thus benefit from a measurement range which extends from 950 hPa to 1040 hPa. In principle, you should not have a need to go beyond that in either direction.

The circuit may be conveniently powered from a 9-volt battery but only if used very occasionally. Since this is usually not the case for a barometer, we advise you to use a mains adaptor instead supplying approximately 9 volts. Calibration basically entails adjusting the potentiometer P1 to light the LED corresponding to the atmospheric pressure of your location at the time. Compare with an existing barometer or, even better, telephone the closest weather station. They will be happy to give you the information. After Evangelista Torricelli, 1608-1647, Italian physician who proved the existence of atmospheric pressure and invented the mercury barometer.
Author: Christian Tavernier - Copyright: Elektor Electronics Magazine

Electronic Torricelli Barometer

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Although it does not have the same charm as real mercury barometers with long glass tubes on pieces of carved and polished wood, the Torricelli barometer discussed here is a functional equivalent and electronic replica of the Torricelli barometer. Actually, rather than displaying the atmospheric pressure on the traditional digital displays, we preferred to reproduce the general look of this respected predecessor of electronic barometers.

The mercury tube is, of course, replaced by a simple LED scale which, if not as beautiful, is still less toxic for the environment in case of breakage. As indicated on the drawing, the pressure sensor utilized is a Motorola MPX2200AP. This circuit is adapted for measuring absolute pressure and has a range well suited for atmospheric pressure. Without entering too deep into the technical details, such sensors deliver an output of voltage proportional not only to the measured pressure but, unfortunately, to their supply voltage as well.

Hence they must be powered from a stable voltage which is ensured here by the use of IC1. Since the output of the MPX2200 is differential and at a very low level, we had to resort to the use of four operational amplifiers IC4.A to IC4.D, contained in one LM324, to obtain levels that can be processed easily. As long as potentiometer P1 is adjusted correctly, this group of operational amplifiers delivers a voltage of 1 volt per atmospheric pressure of 1,000 hPa to the LM3914.


Circuit diagram:
Electronic Torricelli Barometer Circuit Diagram

Since the atmospheric pressure will be within the range 950 to 1040 hPa at sea level, we need to make an expanded-scale voltmeter with this LM3914 in order to better exploit the 10 LEDs that it can control. That is the role of resistors R7 and R8 which artificially raise the minimum voltage value the chip is capable of measuring. Consequently, we can ‘calibrate’ our LED scale with one LED per 10 hPa and thus benefit from a measurement range which extends from 950 hPa to 1040 hPa. In principle, you should not have a need to go beyond that in either direction.

The circuit may be conveniently powered from a 9-volt battery but only if used very occasionally. Since this is usually not the case for a barometer, we advise you to use a mains adaptor instead supplying approximately 9 volts. Calibration basically entails adjusting the potentiometer P1 to light the LED corresponding to the atmospheric pressure of your location at the time. Compare with an existing barometer or, even better, telephone the closest weather station. They will be happy to give you the information. After Evangelista Torricelli, 1608-1647, Italian physician who proved the existence of atmospheric pressure and invented the mercury barometer.
Author: Christian Tavernier - Copyright: Elektor Electronics Magazine
This is a very simple cable TV amplifier using two transistors. This amplifier circuit is most suitable for cable TV systems using 75 Ohm coaxial cables and works fine up to 150MHz. Transistor T1 performs the job of amplification. Up to 20dB gain can be expected from the circuit.T2 is wired as an emitter follower to increase current gain.
assembled on a Vero board.
  • Use 12V DC for powering the circuit.
  • Type no of the transistors are not very critical.
  • Any medium power NPN RF transistors can be used in place of T1 and T2.
  • This is just an elementary circuit. Do not compare it with high quality Cable TV amplifiers available in the market.

Cable TV amplifier

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This is a very simple cable TV amplifier using two transistors. This amplifier circuit is most suitable for cable TV systems using 75 Ohm coaxial cables and works fine up to 150MHz. Transistor T1 performs the job of amplification. Up to 20dB gain can be expected from the circuit.T2 is wired as an emitter follower to increase current gain.
assembled on a Vero board.
  • Use 12V DC for powering the circuit.
  • Type no of the transistors are not very critical.
  • Any medium power NPN RF transistors can be used in place of T1 and T2.
  • This is just an elementary circuit. Do not compare it with high quality Cable TV amplifiers available in the market.
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These days many more audio-visual devices in the home are connected together. This is especially the case with the TV, which may be connected to a DVD player, a hard disk recorder, a surround-sound receiver and often a PC as well. This often creates a problem when earth loops are created in the shielding of the video cables, which may cause hum and other interference. The surround-sound receiver contains a tuner that takes its signal from a central aerial distribution system.

The TV is also connected to this and it’s highly likely that the PC has a TV-card, which again is connected to the same system. On top of this, there are many analogue connections between these devices, such as audio cables. The usual result of this is that there will be a hum in the audio installation, but in some cases you may also see interference on the TV screen.
The ground loop problem can be overcome by galvanically isolating the video connections, for example at the aerial inputs of the surround-sound receiver and the TV.

Special adaptors or filters are sold for this purpose, known as video ground loop isolators. Good news: such a filter can also be easily made at home by yourself. There are two ways in which you can create galvanic isolation in a TV cable. The first is to use an isolating transformer with two separate windings. The other is to use two coupling capacitors in series with the cable. The latter method is easily the simplest to implement and generally works well enough in practice. The simplest way to produce such a ‘filter’ is as an in-line adapter, so you can just plug it onto either end of a TV aerial cable.


Diagram and snapshoot:

Video Isolator Circuit

The only requirements are a male and female coax plug and two capacitors. The latter have to be suitable for high-frequency applications, such as ceramic or MKT types. It is furthermore advisable to choose types rated for high voltages (400 V), since the voltages across these capacitors can be higher than you might expect (A PC that isn’t connected to the mains Earth can have a voltage as high as 115 V (but at a very low, safe current), caused by the filter capacitors in its power supply.

These capacitors don’t need to be high value ones, since they only have to pass through frequencies above about 50 MHz. Values of 1 nF or 2.2 nF are therefore sufficient. To make the isolator you should connect one capacitor between the two earth connections of the coax plugs and the other between the two signal connections. The mechanical construction has to be sturdy enough such that the connections to the capacitors won’t break whenever the inline adapter is removed forcibly.

A good way to do this is to make a cover from a piece of PVC piping for the central part. Wrap aluminium foil round the outside and connect it to one of the plugs, so that the internal parts are properly shielded from external interference. Make sure that the aluminium foil doesn’t make contact with the other plug, otherwise you lose the isolation. The majority of earth loops will disappear when you connect these filters to all used outputs of the central aerial distribution system where the signal enters the house.
Harry Baggen
Elektor Electronics 2008

Video Isolator Circuit Diagram

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These days many more audio-visual devices in the home are connected together. This is especially the case with the TV, which may be connected to a DVD player, a hard disk recorder, a surround-sound receiver and often a PC as well. This often creates a problem when earth loops are created in the shielding of the video cables, which may cause hum and other interference. The surround-sound receiver contains a tuner that takes its signal from a central aerial distribution system.

The TV is also connected to this and it’s highly likely that the PC has a TV-card, which again is connected to the same system. On top of this, there are many analogue connections between these devices, such as audio cables. The usual result of this is that there will be a hum in the audio installation, but in some cases you may also see interference on the TV screen.
The ground loop problem can be overcome by galvanically isolating the video connections, for example at the aerial inputs of the surround-sound receiver and the TV.

Special adaptors or filters are sold for this purpose, known as video ground loop isolators. Good news: such a filter can also be easily made at home by yourself. There are two ways in which you can create galvanic isolation in a TV cable. The first is to use an isolating transformer with two separate windings. The other is to use two coupling capacitors in series with the cable. The latter method is easily the simplest to implement and generally works well enough in practice. The simplest way to produce such a ‘filter’ is as an in-line adapter, so you can just plug it onto either end of a TV aerial cable.


Diagram and snapshoot:

Video Isolator Circuit

The only requirements are a male and female coax plug and two capacitors. The latter have to be suitable for high-frequency applications, such as ceramic or MKT types. It is furthermore advisable to choose types rated for high voltages (400 V), since the voltages across these capacitors can be higher than you might expect (A PC that isn’t connected to the mains Earth can have a voltage as high as 115 V (but at a very low, safe current), caused by the filter capacitors in its power supply.

These capacitors don’t need to be high value ones, since they only have to pass through frequencies above about 50 MHz. Values of 1 nF or 2.2 nF are therefore sufficient. To make the isolator you should connect one capacitor between the two earth connections of the coax plugs and the other between the two signal connections. The mechanical construction has to be sturdy enough such that the connections to the capacitors won’t break whenever the inline adapter is removed forcibly.

A good way to do this is to make a cover from a piece of PVC piping for the central part. Wrap aluminium foil round the outside and connect it to one of the plugs, so that the internal parts are properly shielded from external interference. Make sure that the aluminium foil doesn’t make contact with the other plug, otherwise you lose the isolation. The majority of earth loops will disappear when you connect these filters to all used outputs of the central aerial distribution system where the signal enters the house.
Harry Baggen
Elektor Electronics 2008
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